Near-end, far-end and echo cancellers in a multi-channel transceiver system

ABSTRACT

A transceiver according to the present invention receives data from a plurality of frequency separated transmission channels from a complementary transmitter and includes an interference filter for correcting for interference from transmitters other than the complementary transmitter. The interference filter, for example, can correct for near-end cross-talk and echo interference filtering and/or far-end crosstalk interference filtering is presented. A transceiver can include a transmitter portion and a receiver portion with one or more receivers coupled to receives signals in the plurality of frequency separated transmission channels. A baseband transmitter can be combined with one or more transmitters that transmit data into one of the frequency separated transmission bands. Any combination of modulation systems can be utilized (e.g. PAM for the baseband and QAM for the frequency separated bands). In some embodiments, one baseband PAM transmitter is combined with one or more frequency separated QAM transmitters.

This is a continuation of application Ser. No. 10/454,382, filed Jun. 3,2003 now U.S. Pat. No. 7,388,904, which is incorporated herein byreference.

BACKGROUND

1. Field of the Invention

The present invention is related to high-speed communications of data ina communication system and, in particular, to high data ratetransmission of data between components in a communication system.

2. Discussion of Related Art

There is currently a great deal of interest in high speed transceiversystems, both for communications in an intranet environment and forcommunications between components in various systems. As an example of ahigh data rate system, high-speed Ethernet local area networks (LANs),100 BASE-TX Ethernet and 1000 Base TX Ethernet (1 Gigabit/s) usingcategory-5, 5E or 6 copper wire, are being developed. These high speedsystems require new techniques in high-speed data processing. High-speeddata transmission techniques are also useful in wide-area networks anddigital subscriber loop applications. High data rate transceiver systemsare also utilized in many back-plane environments, including opticalswitching devices, router systems, switches, and storage area networkingswitches. Other environments that utilize high speed communicationbetween components include inter-cabinet communications and chip-to-chipcommunications.

Typically, data is transferred in a communication system by transmittingsignals having voltages from sets of voltages referred to as symbolsets. Each symbol (i.e. voltage level) in the symbol set represents oneor more digital bits of data. Existing techniques utilized in suchenvironments typically use non-return to zero (NRZ) modulation to sendand receive information over high-bandwidth transmission media. Othercommon symbol sets include MLT3, PAM or QAM systems. Typically, thetransceiver for sending high-speed data over such networks is called aserializer/deserializer, or SERDES, device.

FIG. 1A shows a block diagram of a typical transceiver environment 100.Components 101-1 through 101-Q are coupled to transmit and receive datathrough input/output (I/O) ports, 102-1 through 102-Q, respectively,which are coupled through transmission medium 110. Conventionally,components 101-1 through 101-Q are SERDES devices. Transceiverenvironment 100 can represent either a back-plane environment (wherecomponents 101-1 through 101-Q are physically relatively close to oneanother) or a networking environment (where components 101-1 through101-Q are more separated).

FIG. 1B shows a block diagram of a conventional transmitter portion ofone of SERDES devices 101-1 through 101-Q on I/O ports 102-1 through102-Q, respectively. Parallel data is received in a bit encoder 105. Bitencoder 105 encodes the parallel data, for example by adding redundancyin the input data, to ensure a minimum rate of data transitions in theoutput data stream. Typical encoding schemes include rate 8/10 (8 bitinput to 10 bit output) encoding. The parallel data is serialized inparallel to serial converter 106. Output driver 107 then receives theserialized data from parallel to serial converter 106 and outputs,usually, a differential voltage signal for transmission overtransmission medium 110. In addition, there is typically a phase lockedloop (PLL) 114 that provides the necessary clock signals for encoder 105and parallel-to-serial converter 106. The input signal to PLL 114 is areference clock signal from a system PLL 103.

FIG. 1C shows an example of a conventional receiver 108 of one of SERDESdevices 101-1 through 101-Q on I/O ports 102-1 through 102-Q,respectively, of FIG. 1A. Input driver 109 receives a differentialvoltage signal from transmission medium 110 and outputs the analog datasignal to clock and data recovery circuit 115. Data recovery 115 can, insome systems, perform equalization, recover the timing, and output aserial bit stream of data to serial-to-parallel converter 111. Theserial data is input to bit decoder 112, which converts the paralleldata to parallel decoded data. Clock and data recovery circuit 115 alsooutputs the necessary clock signals to serial-to-parallel converter 111and bit decoder 112.

The actual demands for the various data transmission environments mayvary widely (e.g., LAN environments have different transmissionrequirements from back-plane environments). A conventional SERDES system100 for a back-plane environment, for example, can enable serial datacommunication at data rates as high as 2.5 Gbps to 3.125 Gbps over apair of FR4 copper traces in a copper back-plane communication system.Current systems utilizing category 5, 5E or 6 copper wire can enableserial data communications rates as high as 1 Gbit/sec using GigabitEthernet. One of the biggest problems with existing SERDES systems 100is that they are very bandwidth inefficient, i.e., they require 3.125GHz of bandwidth to transmit and receive 2.5 Gbps of data over a singlepair of copper wires. Therefore, it is very difficult to increase thedata rates across bus 110. Additionally, SERDES system 100 requires theimplementation of a high clock rate (3.125 GHz for 2.5 Gbps data rates)phase locked loop (PLL) 114 implemented to transmit data and recoverhigh clock rates in data recovery 115. The timing window within whichreceiver 108 needs to determine whether the received symbol in datarecovery 115 is a 1 or a 0 is about 320 ps for the higher data ratesystems. This timing window creates extremely stringent requirements onthe design of data recovery 115 and PLL 114, as they must have very lowpeak-to-peak jitter.

Conventional networking environments operate at slower baud rates, butsuffer from similar difficulties. As an example, a 1 Gigabit transfercan be accomplished through transmitting PAM-5 data at 125 MHz throughfour (4) twisted copper pair. It would be desirable to allow higher datarates in networking environments.

Conventional SERDES system 100 also suffers from other problems,including eye closure due to intersymbol interference (ISI) from thedispersion introduced by transmission medium 110. The ISI is a directresult of the fact that the copper traces of transmission medium 110attenuate higher frequency components in the transmitted signals morethan the lower frequency components in the transmitted signals.Therefore, the higher the data rate the more ISI suffered by thetransmitted data. In addition, electrical connectors and electricalconnections (e.g., vias and other components) used in SERDES device 100cause reflections, which also cause ISI.

To overcome these problems, equalization must be performed on thereceived signal in data recovery 115. However, in existing very highdata-rate communication systems, equalization is very difficult toperform, if not impossible due to the high baud rate. A more commonlyutilized technique for combating ISI is known as “pre-emphasis”, orpre-equalization, performed in bit encoder 105 and output driver 107during transmission. In some conventional systems, the amplitude of thelow-frequencies in the transmitted signal is attenuated to compensatefor the higher attenuation of the high frequency component by thetransmission medium of bus 110. While this makes the receiver morerobust to ISI, pre-emphasis reduces the overall noise tolerance oftransmission over transmission medium 110 of communication system 100due to the loss of signal-to-noise ratio (SNR). At higher data rates,conventional systems quickly become intractable due to the increaseddemands.

Another difficulty with conventional SERDES system 100 is withcorrection for near-end cross talk (NEXT) interference, Far-end crosstalk interference (FEXT) and echo cancellation. NEXT interferencerelates to the interference between the transmitter portions of device101-q, an arbitrary one of devices 101-1 through 101-Q, and the receiverportions of device 101-q, where the interfering transmitter portion istransmitting on a separate conductor from the receiver portion. Echorefers to the interference between the transmitter portions of device101-q and the receiver portions of device 101-q, where the transmitterportion is transmitting on the same wire as the receiver portion. FEXTrefers to interference between transmitters of counterpart transmitterportions transmitting to the receiver portion of device 101-q. In manycases, transmitter 104 and receiver 108 of device 101-q are adjacentlylocated as transceiver SERDES device 101-1 and may share a bus line tobus 110. Further, in many cases device 101-q is in communication withone or more counterpart ones of devices 101-1 through 101-Q that canprovide interference to the receiver portion of device 101-q. Forexample, NEXT, FEXT and Echo interference can be problematic in systemsutilizing category 5, 5E or 6 cabling, which are capable of supportinghigh transmission rates.

Therefore, there is a need for a more robust system for transmittingdata in a transmission system at very high speeds.

SUMMARY

In accordance with the present invention, a transceiver is presentedthat allows very high data transmission rates over a data bus thatutilizes the signal attenuation properties of the interconnect systemand which corrects for interference at a receiver from the effects oftransmitters other than the complementary transmitter to the receiver.This interference can include near-end cross talk (NEXT), far-end crosstalk (FEXT) and/or Echo interference.

A transceiver can include one or more individual transmitters and one ormore individual receivers, where at least some of the transmitterstransmit data on a plurality of frequency separated channels. Atransceiver according to the present invention includes an interferencefilter, which can include a NEXT, FEXT, and/or Echo filter, thatcorrects transmitted data received by each of the one or more receiversin the transceiver from interference generated by the one or moreadjacent transmitters in the transceiver or from one or more adjacenttransmitters to the complementary transmitter to the receivers of thetransceiver.

Therefore, a transceiver according to the present invention includes areceiver portion including at least one receiver to receive signals froma complementary transmitter through a transmission medium, the at leastone receiver including a plurality of demodulators to receive thesignals from a corresponding plurality of frequency separated channels;and an interference filter coupled to the receiver portion tosubstantially reduce interference in signals received by the receiverportion that result from transmission coupled to each of the pluralityof corresponding frequency separated channels from transmitters otherthan the complementary transmitter. The interference filter can includeany number of filter, including a far-end cross-talk (FEXT) filter and anear-end cross-talk and echo filter.

A transmitter in accordance with the present invention can include anynumber of transmitters and at least one receiver. Each of thetransmitters can be in communications with a complementary receiver of aseparate transceiver. Additionally, each of the at least one receiverscan be in communications with a complementary transmitter of a separatetransceiver. Each receiver, then, is in communications with acomplementary transmitter of a different transceiver and is adjacent toone or more transmitters with which it forms the transceiver. In somecases, all the transmitters and receivers from one transceiver willcommunicate with corresponding counterparts of a single far endtransceiver.

In the data transmission system, a transmitter from a first transceiveris coupled with a receiver from a second transceiver through atransmission medium. The transmitter receives parallel data having Nbits and separates the N bits into subsets for transmission. In someembodiments, the N-bits are separated into (K+1) subsets fortransmission into the base band and K frequency separated channels. Insome embodiments, the N-bits are separated into K subsets fortransmission into K frequency separated channels. The transmitter iscoupled to transmit signals on the transmission medium. The K subsets ofdata for transmission into the K frequency separated channels areup-converted to frequencies corresponding to those channels. The summedoutput signal resulting from the summation of the K up-convertedchannels and the base-band channel, if present, is transmitted over thetransmission medium.

A receiver of the transceiver receives data from a complementarytransmitter through the transmission medium. In some embodiments, datafrom the base-band and the K frequency separated channels from thetransmission medium is received and the parallel bits of datatransmitted by the complementary transmitter is recovered. In someembodiments, data from K separated channels is received from thetransmission medium and the base-band channel is not utilized.

In addition to interference caused by near-end cross talk from adjacenttransmitters, far end cross talk from other far end transmitters, andecho from a transmitter on the same conducting media (in the case ofCat-5, 5E or 6 cable or copper pair) to data received by a receiver in atransceiver according to the present invention, the data can furthersuffer from inter-symbol interference (ISI) as well as cross-channelinterference. Cross-channel interference is due to harmonic generationin up-conversion and down-conversion processes between the communicatingtransmitter and receiver pair. Therefore, embodiments of the presentinvention can also include filters to address other interferencemechanisms, for example intersymbol interference and cross-channelinterference in the data.

These and other embodiments are further discussed below with respect tothe following figures.

BRIEF DESCRIPTION OF THE FIGURES

FIGS. 1A, 1B and 1C show block diagrams for a conventional system oftransmitting data over a back-plane.

FIG. 2A shows a block diagram of a transmission system according to thepresent invention.

FIG. 2B shows a block diagram of a transceiver according to the presentinvention.

FIG. 2C shows a block diagram of a transmitter of the transceiver shownin FIG. 2B according to the present invention.

FIG. 2D shows a block diagram of a receiver of the transceiver shown inFIG. 2B according to the present invention.

FIG. 2E shows a block diagram of an embodiment of a transceiver pairconfiguration according to the present invention.

FIG. 2F shows a block diagram of another embodiment of a transceiverpair configuration according to the present invention.

FIG. 3 shows a graph of attenuation versus transmission band on thetransmission medium according to the present invention.

FIGS. 4A and 4B show block diagrams of embodiments of transmissionmodulators according to the present invention.

FIG. 5 shows a block diagram of an embodiment of a receiver according tothe present invention.

FIG. 6 shows a block diagram of a portion of an embodiment of a near-endcross talk (NEXT) interference filter according to the presentinvention.

FIG. 7 shows a block diagram for a portion of an embodiment of a far-endcross talk (FEXT) interference filter according to the presentinvention.

In the figures, elements designated with the same identifications onseparate figures are considered to have the same or similar functions.

DETAILED DESCRIPTION

FIG. 2A shows a block diagram of a transmission system 200 according tothe present invention. System 200 includes any number of components201-1 through 201-P, with component 201-p representing an arbitrary oneof components 201-1 through 201-P, coupled through a transmission medium110. Transmission medium 110 may couple component 201-p to all of thecomponents 201-1 through 201-P or may couple component 201-p to selectedones of components 201-1 through 201-P. In some embodiments, individualtransmitters and receivers of components 201-1 through 201-P are coupledtogether through category 5, 5E or 6 twisted copper pair. Further, insome embodiments, transmission medium 110 can include a router to form atransmission network, for example a hub and spoke network, fortransmission of data between individual ones of components 201-1 through201-P.

System 200 can represent any transmission system, for example a localarea network (LAN), wide area network (WAN), digital subscriber loop,chassis-to-chassis digital communication system, or chip-to-chipinterconnect with components 201-1 through 201-P representing individualcomputer systems, cards, cabinets, or chips.

Transmission channel 110 can represent any transmission channel,including optical channels, infrared channels, wireless channels,multiple twisted copper pair (such as Category-5, 5E or 6 cable), copperwire, FR4 copper traces, or copper based back-plane interconnectchannel. Additionally, any conducting medium can be utilized intransmission channel 110. Transmission channel 110 may further includenetworking devices such as routers to direct connections betweenindividual components. Typically, transmission channel 110 attenuateshigher frequency signals more than lower frequency signals. As a result,intersymbol interference problems are typically greater for high datarate transmissions than for low data rate transmissions. In addition,cross-talk from neighboring signals increases with transmissionfrequency.

As an example, a transmission system utilizing multiple pairs of twistedcopper pair is discussed in this disclosure. It should be noted thatother transmission media in other transmission environments can be usedfor the transmission system.

Components 201-1 through 201-P include transceivers 255-1 through 255-P,respectively. Each of transceivers 255-1 through 255-P, in turn,includes transmitter portions 210-1 through 210-P, respectively, andreceiver portions 220-1 through 220-P, respectively. Each of transmitterportions 210-1 through 210-P includes one or more individualtransmitters according to the present invention and each of receiverportions 220-1 through 220-P includes one or more individual receiversaccording to the present invention. In some embodiments, certain ones ofcomponents 201-1 through 201-P may only include a transmitter portionand others may include only a receiver portion. Therefore, in someembodiments of transmission system 200, some of transmitter portions210-1 through 210-P may be absent and some of receiver portions 220-1through 220-P may be absent.

FIG. 2B shows an embodiment of transceiver 255-p, an arbitrary one oftransceivers 255-1 through 255-P, according to the present invention.The embodiment shown in FIG. 2B includes filters for reducing theinterference in signals received by receiver portion 220-p fromtransmitters other than those involved in transmitting the signalsinterfering with those signals. In FIG. 2B, NEXT and Echo filters 250 aswell as a FEXT filter 251 are shown. Generally, one or all of thefilters may be present in an embodiment of transceiver 255-p accordingto the present invention.

Transmitter portion 210-p of transceiver 255-p includes transmitters270-1 through 270-T, with transmitter 270-t indicating an arbitrary oneof transmitters 270-1 through 270-T. Transmitter 270-t receives N_(t)^(T) bits from data allocation 271 and transmits data corresponding withthe N_(t) ^(T) bits over transmission media 110. As shown in FIG. 2B,data from transmitter 270-t in the embodiment shown is transmitted overa pair of wires.

Receiver portion 220-p of transceiver 255-p includes receivers 272-1through 272-R, with receiver 272-r indicating an arbitrary one ofreceivers 272-1 through 272-R. Receiver 272-r receives data fromtransmission medium 110 and outputs N_(r) ^(R) bits to data parsing 273.Data parsing 273 outputs a data stream received by receiver portion220-p and a receive clock corresponding to the data stream. Note thatthe number of transmitters, T, and the number of receivers, R, intransceiver 255-p need not be the same.

In some embodiments of the invention, a single copper pair oftransmission medium 110 can be utilized to both transmit and receive.For example, transmitter 270-t and receiver 272-r may share a singlepair of copper wire. This shared-wire configuration results in echointerference between transmitter 270-t and receiver 272-r. Echointerference refers to the receipt of the transmitted signal fromtransmitter 270-t at receiver 272-r in the shared-wire configurationfrom two sources: The transmitted signal from transmitter 270-t appearsdirectly on receiver 272-r; and the reflection of the transmitted signalfrom transmitter 270-t by the corresponding receiver to transmitter270-t (i.e., the receiver to which transmitter 270-t is sending data)and by any other connections and impedance mismatches in transmissionmedium 110 that would reflect the transmitted signal back to be detectedby receiver 272-r.

In addition to echo interference at receiver 272-r, each of transmitters270-1 through 270-T may be closely enough spaced to receiver 272-r suchthat signals transmitted by each of transmitters 270-1 through 270-T isreceived by receiver 272-r. Signals transmitted by transmitters 270-1through 270-T can also be received by receiver 272-r through theproximity of wires in transmission medium 110, for example because ofsignals from one copper pair leaking to another copper pair.Interference by receipt of signals from transmitters 270-1 through 270-Tat receiver 272-r, or NEXT interference, therefore, may also be present.

Therefore, to correct for Next and echo interference, some embodimentsof transceiver 255-p can include a NEXT/Echo filter 250. NEXT/Echofilter 250 receives input data from each of transmitters 270-1 through270-T and outputs correction data to each of receivers 272-1 through272-R. In some embodiments, portions of NEXT/Echo filter 250 may bedistributed throughout each of receivers 272-1 through 272-R. Next/Echofilter 250 corrects each of receivers 272-1 through 272-R for echointerference and NEXT interference caused by transmitters 270-1 through270-T, especially if transmitter/receiver pairs share transmissionmedia.

Receivers 272-1 through 272-R may also include a FEXT filter 251. FEXTfilter 251 receives inputs from each of receivers 272-1 through 272-Rand outputs correction data to each of receivers 272-1 through 272-R.FEXT filter 251, therefore, corrects for cross interference betweentransmitters corresponding to each of receivers 272-1 through 272-R(i.e., each of the transmitters that are transmitting data to receivers272-1 through 272-R).

In operation, one or more of transmitters 270-1 through 270-T fromcomponent 201-p is in communication with complementary receivers inothers of components 201-1 through 201-P. Additionally, one or more ofreceivers 272-1 through 272-R can be in communication with complementarytransmitters in others of components 201-1 through 201-P. Further, insome embodiments, timing for all of components 201-1 through 201-P canbe provided by a phase-locked-loop (PLL) 203 synchronized to a transmitsource clock signal. In some embodiments, PLL 203 provides a referenceclock signal and each of components 201-1 through 201-P can include anynumber of phase locked loops to provide internal timing signals.

In some embodiments, each of components 201-1 through 201-P will haveits own reference clock, which is compensated with a frequencyadjustment circuit. As discussed in U.S. patent application Ser. No.10/410,255, filed on Dec. 4, 2002, and U.S. patent application Ser. No.10/167,158, filed on Jun. 10, 2002, each of which is herein incorporatedby reference in their entirety, the timing between components 201-1through 201-P is matched such that the base-band frequencies are thesame for each of components 201-1 through 201-P and the up-conversionand down-conversion mixers between complementary transmitter/receiverpairs operate at the same frequency. The complexity of systems whereeach of components 201-1 through 201-P operate at different frequenciesmay be highly increased.

In some systems, for example back-plane systems or cabinetinterconnects, the transmission distance through transmission channel110, i.e., the physical separation between components 201-1 through201-P, can be as low as 1 to 1.5 meters. In some chip-to-chipenvironments, the physical separation between components 201-1 though201-P can be much less (for example a few millimeters or a fewcentimeters). In local area network or wide area network applications,separations between components 201-1 through 201-P can be up to 100 mfor LAN and several kilometers for WAN applications. Furthermore, insome embodiments transmission channel 110 can be multiple twisted copperpair (or any other current carrying wire configuration) carryingdifferential signals between components 201-1 through 201-P, for examplecategory 5, 5E or 6 cabling. In some embodiments, components 201-1through 201-P can share wires so that fewer wires can be utilized. Insome embodiments, however, dedicated conducting paths can be coupledbetween at least some of components 201-1 through 201-P. Further,transmission medium 110 can be an optical medium, wireless medium, ordata bus medium.

Each of transmitters 270-1 through 270-T and receivers 272-1 through272-R of transceiver 255-p can be in communication with complementarytransmitters and receivers from one or more transceivers of components201-1 through 201-P. For example, each of transmitters 270-1 through270-T may be in communication with complementary receivers of one ormore of components 201-1 through 201-P. Furthermore, each of receivers272-1 through 272-R is in communication with complementary transmittersfrom one or more of components 201-1 through 201-P, but not necessarilyall from the same one of components 201-1 through 201-P. It should benoted that transmitter 210-p and receiver 220-p can communicateseparately with any combination of receivers and transmitters,respectively, of transceivers 255-1 through 255-P of components 201-1through 201-P, respectively. In the particular embodiments discussed inthis disclosure, each of components 201-1 through 201-P are incommunications with a complementary counterpart one of components 201-1through 201-P.

FIG. 2C shows a block diagram of an embodiment of transmitter 270-t, anarbitrary one of transmitters 270-1 through 270-T included intransceiver 255-p, according to the present invention. Transceiver 255-pis an arbitrary one of transceivers 255-1 through 255-P. For ease ofdiscussion, the subscripts on designations of individual elementsindicating particular transceiver and which transmitter or receiver inthe particular transceiver may be neglected, but will be added if itclarifies the discussion. The particular transmitter or receiver andtransceiver is clear in context and discussions with regard to aparticular transmitter or receiver component can be extended to othertransmitters and receivers.

Transmitter 270-t receives an N_(t) ^(T)-bit parallel data signal at abit allocation block 211 to be transmitted over media 110. Bitallocation block 211 also receives the reference clock signal from PLL203. Bit allocation block 211 segregates the N_(t) ^(T) input bits intogroups of bits allocated to the multiple channels. In the embodimentshown in FIG. 2C, allocation block 211 segregates the N_(t) ^(T) inputbits into K+1 individual channels such that there are n₁ through n_(K)bits input to up-converting modulators 212-1 through 212-K,respectively, and n₀ bits input to base-band modulator 217. Someembodiments do not include base-band modulator 217. Base-band modulator217 and up-converting modulators 212-1 through 212-K, collectively,transmit into (K+1) channels. In some embodiments, each of the N_(t)^(T) bits is assigned to one of the K+1 individual channels so that thesum of n₀ through n_(K) is the total number of bits N_(t) ^(T). In someembodiments, bit allocation block 211 may include error coding,redundancy, or other overall encoding such that the number of bitsoutput by bit allocation block 211, i.e.

${\sum\limits_{i = 0}^{K}n_{i}},$is greater than N_(t) ^(T).

Each of up-converting modulators 212-1 through 212-K encodes the digitaldata input to it and outputs a signal modulated at a different carrierfrequency. Therefore, the n_(k) digital data bits input to up-convertingmodulator 212-k, an arbitrary one of up-converting modulators 212-1through 212-K of transmitter 270-t, is output as an analog signal in akth transmission channel at a carrier frequency f_(k). Additionally,base-band modulator 217, if present, transmits into the base-bandchannel. A discussion of embodiments of transmitter 270-t is furtherincluded in U.S. application Ser. No. 09/904,432, filed on Jul. 11,2001, U.S. application Ser. No. 09/965,242, filed on Sep. 26, 2001,application Ser. No. 10/071,771, filed on Feb. 6, 2002, and applicationSer. No. 10/310,255, filed on Dec. 4, 2002, each of which is assigned tothe same assignee as is the present disclosure and is hereinincorporated by reference in its entirety.

FIG. 3 shows schematically the transport function for a typicaltransmission channel, H(f), of transmission medium 110. As is shown, theattenuation at higher frequencies is greater than the attenuation atlower frequencies. Up-converting modulators 212-1 through 212-K transmitanalog data at carrier frequencies centered about frequencies f₁ throughf_(K), respectively. Therefore, modulators 212-1 through 212-K transmitinto transmission channels 301-1 through 301-K, respectively. Base-bandmodulator 217 transmits into transmission channel 301-0, which iscentered at 0 frequency. In some embodiments, the width of each oftransmission channels 301-0 through 301-K can be the same. The width ofthe bands of each of transmission channels 301-0 through 301-K can benarrow enough so that there is little to no overlap between adjacentones of transmission channels 301-0 through 301-K. In some embodiments,since the attenuation for the lower frequency channels is much smallerthan the attenuation for the higher frequency channels, lower frequencychannels can be bit-loaded to carry a higher number of bits per baudinterval than the number of bits per baud interval that can be carriedat higher carrier frequencies.

As shown in FIG. 2C, the analog output signal from each of up-convertingmodulators 212-1 through 212-K, y₁(t) through y_(K)(t), then representsthe transmission signal in each of channels 301-1 through 301-K,respectively. Signals y₁(t) through y_(K)(t), then, are input to summer213 and the summed analog signal output from summer 213 can be input toa high pass filter 215. The output signal from high pass filter 215 isinput to summer 216 where it is summed with the base-band signal y₀(t)from base-band modulator 217. High pass filter 215 preventsup-converting modulators 212-1 through 212-K from transmitting signalsinto the base-band channel and reduces or eliminates the need toconsider cross-channel interference between signals produced bybase-band modulator 217 and those generated by up-converting modulators212-1 through 212-K.

The output signal from summer 216, z(t), is input to an output driver214. In some embodiments, output driver 214 generates a differentialtransmit signal corresponding to signal z(t) for transmission overtransmission medium 110. Output driver 214, if transmission medium 110is an optical medium, can also be an optical driver modulating theintensity of an optical signal in response to the signal z(t). Thesignal z(t), after transmission through transmission medium 110, isreceived by a complementary receiver in one of components 201-1 through201-P.

FIG. 2D shows an example of a receiver 272-r of receiver portion 220-pof transceiver 255-p. Receiver 272-r can receive a differential receivesignal, which originated from a complementary transmitter from anotherof components 201-1 through 201-P, into an input buffer 224. In someembodiments, an optical signal can be received at input buffer 224, inwhich case input buffer 224 includes an optical detector. The outputsignal from input buffer 224, Z(t), is closely related to the outputsignal from the complementary transmitter, but shows the effects oftransmission through transmission medium 110, including intersymbolinterference (ISI). Additionally, near-end crosstalk interference andpossibly Echo interference from the signals transmitted by transmitter210-p of transceiver 255-p will also be included in signal Z(t). Also,the signal Z(t) may include FEXT interference resulting from signalstransmitted by transmitters adjacent to the complementary transmitter.

The signal Z(t) is input to each of down-converting demodulators 222-1through 222-K and into base-band demodulator 223. Down-convertingdemodulators 222-1 through 222-K demodulate the signals from each of thetransmission channels 301-1 through 301-K, respectively, and recoversthe bit stream from each of carrier frequencies f₁ through f_(K),respectively. Base-band demodulator 223 recovers the bit stream whichhas been transmitted into the base-band channel, if the base-bandchannel is present. The output signals from each of down-convertingdemodulators 222-1 through 222-K, then, include n₁ through n_(K)parallel bits, respectively, and the output signal from base-banddemodulator 223 include n₀ parallel bits. In the embodiment shown inFIG. 2D, each of base-band demodulator 223 and down-convertingdemodulators 222-1 through 222-K can be coupled to Next/Echo filter 250,can be coupled to FEXT filter 251, and can be coupled to receive signalsfrom each of the others of base-band demodulator 223 and down-convertingdemodulators 222-1 through 222-K. Therefore, in some embodiments each ofbase-band demodulator 223 and down-converting demodulators 222-1 through222-K can correct for NEXT and Echo interference, FEXT interference, andcross-channel coupling interference. As discussed above, embodiments ofreceiver 272-r according to the present invention can include one ormore of Next/Echo filter 250 and FEXT filter 251.

Although the examples discussed here describe a transmitter with (K+1)channels and a receiver with (K+1) channels, one skilled in the art willrecognize that transmitter 270-t of transceiver 255-p shown in FIG. 2Cand receiver 272-r of transceiver 255-p may use different numbers oftransmission channels centered upon different transmission frequenciesf₁ through f_(K). A complementary transceiver/receiver pair (i.e.,transmitter 270-t and the receiver coupled to transmitter 270-t throughtransmission medium 110, or receiver 272-r and the transmitter coupledto receiver 272-r through transmission medium 110) utilizes a common setof transmission channels. In the embodiments specifically describedbelow, the transmission frequencies f₁ through f_(K) for each oftransceivers 255-1 through 255-P are integral multiples of a frequencyf₀.

As shown in FIG. 2D, the output signals from base-band demodulator 223and down-converting demodulators 222-1 through 222-K are input to bitparsing 221 where the transmitted signal having N_(r) ^(R) parallel bitsis reconstructed. Receiver 272-r also receives the reference clocksignal from PLL 203, which can be used to generate internal timingsignals. Furthermore, receiver system 220-p outputs a receive clocksignal with the N-bit output signal from bit parsing 221. In someembodiments, each of transceivers 255-1 through 255-P includes timingrecovery in order to match the data transmission timing betweencomplementary transmitter/receiver pairs. In embodiments where each oftransmitters 270-1 through 270-T is in communication with complementaryreceivers of another transceiver and receivers 272-1 through 272-R arein communication with complementary transmitters of the same othertransceiver, timing can be matched between transceiver pairs.

Down-converting demodulators 222-1 through 222-K, and in someembodiments base-band demodulator 223, can be further coupled so thatcross-channel interference can be cancelled. In embodiments where filter215 of transmitter 210-p is not present or does not completely removethe base-band from the output signal of adder 213, then cross-channelinterference in the base-band channel also may need to be considered.Due to the mixers in the up-conversion process, multiple harmonics ofeach signal may be generated from each modulator in the complementarytransmitter. For example, using an embodiment of transmitter 270-t as anexample, up-converting modulators 212-1 through 212-K can transmit atcarrier frequencies f₁ through f_(K) equal to f₀, 2f₀ . . . Kf₀,respectively. Base-band modulator 217 transmits at the base-bandfrequency, e.g. base-band modulator 217 transmits with no carrier.

Due to the harmonics in the mixer of transmitter 272-r, the signaltransmitted at carrier frequency f₁ will also be transmitted in the baseband and at frequencies 2f₁, 3f₁, . . . . Additionally, the signaltransmitted at carrier frequency f₂ will also be transmitted in the baseband and at 2f₂, 3f₂, . . . . Therefore, any time any of the bandwidthof any harmonics of the channels overlap with other channels or theother channel's harmonics, significant cross-channel symbol interferencecan occur due to harmonics in the mixers of up-converting modulators212-1 through 212-K. For example, in the case where the carrierfrequencies are multiples of f₀, channel 1 transmitting at f₀ will alsotransmit at 0, 2f₀, 3f₀, . . . , i.e. into each of the other channels.

Similarly, the complementary transmitter to receiver 272-r will generatecross-channel interference. Additionally, the down converters ofdown-converting demodulators 222-1 through 222-K of receiver 272-r alsocreate harmonics, which means that some of the transmission of the thirdchannel will be down-converted into the first channel, for example.Therefore, further cross-channel interference can be generated in thedown-conversion process of receivers 221-1 through 221-K of transceiver255-p. Some embodiments of the present invention correct for thecross-channel symbol interference as well as the inter-symbolinterference. Note that it is well known that if the duty cycle of theharmonic wave that is being mixed with an input signal is 50%, only oddharmonics will be generated. Even harmonics require higher or lower dutycycles.

In some embodiments, the symbol baud rates for each of channels 301-0through 301-K can be the same. In some embodiments, bit-loading can beaccomplished by varying the symbol sets for lower frequency componentssuch that a higher number of bits can be encoded.

As shown in FIGS. 2B and 2C, a signal from each of base-band modulator217 and up-converting modulators 212-1 through 212-K from each oftransmitters 270-1 through 270-T are input to NEXT/Echo filter 250.NEXT/Echo filter 250, which in some embodiments can be distributedthrough base-band demodulator 223 and down-converting demodulators 222-1through 222-K of each of receivers 272-1 through 272-R, calculatescorrections to the data received by each of receivers 272-1 through272-R. The data received in each of base-band demodulators 223 anddown-converting demodulators 222-1 through 222-K of receiver 272-r, forexample, is corrected by NEXT/Echo filter 250 for near-end cross talkand echo interference.

For many of the same reasons discussed above with respect to across-channel interference filter, each of base-band modulator 217 andup-converting modulators 212-1 through 212-K may interfere with each ofbase-band demodulator 223 and down-converting demodulators 222-1 through222-K, even if they operate at different frequencies. Due to theharmonics, interference at one modulator frequency may interfere withreceive signals at different demodulator frequencies. In embodiments ofthe invention where one of transmitters 270-1 through 270-T shares asingle connection in transmission medium 110 with one of receivers 272-1through 272-R, then correction for echo interference may also beaccomplished by NEXT/Echo filter 250. Again, due to the harmonics, eachof base-band demodulator 223 and down-converting demodulators 222-1through 222-K can be corrected for each of base-band modulators 217 andup-converting modulators 212-1 through 212-K of a transmitter 270-twhich shares the connection.

Additionally, FEXT filter 251 corrects for interference fromtransmitters adjacent to the transmitter coupled to receiver 272-r. Formany of the same reasons as above, other transmitters which are adjacentto the complementary transmitter of receiver 272-r interfere withsignals received by each of base-band demodulator 223 anddown-converting demodulators 222-1 through 222-K of receiver 272-r.

FIG. 4A shows an embodiment of base-band modulator 217 of transmitter270-t (FIG. 2C). Base-band modulator 217 may include a scrambler 454 andencoder 455. Scrambler 454 functions to whiten the data. Encoder 455encodes the n₀ bits input to base-band modulator 217 to n₀+l bits. Theoutput signal from encoder 455 is then input to symbol mapper 456.Symbol mapper 456 converts the n₀+l parallel bits into a symbol fortransmission. In some embodiments, symbol mapper 456 can be a PAMencoder. The PAM symbol set can be of any size. In some embodiments, forexample, a 16 level symbol set (16-PAM) can be utilized to representn₀+l=4 parallel bits. Encoder 455 can provide 3/4 encoding or noencoding. The output signal from symbol mapper 456 is input todigital-to-analog converter 457 which converts the symbol set determinedby symbol mapper 456 into the corresponding output voltages.

In some embodiments, the analog output signal from DAC 457 isprefiltered through filter 458. In some embodiments, filter 458 mayprepare the output signal for transmission through medium 110 (see FIG.2A) so that the signal received by a receiver is corrected fordistortions caused by the channel. For example, if the base-band channelof transmission medium 110 is known to have a transfer function of(1+D(z)), then filter 458 may execute a transfer function equal to1/(1+D(z)) in order to cancel the transfer function of transmissionmedium 110. The output signal from filter 458 can be input to low-passfilter 459. Filter 459 removes the higher frequency content, which mayinterfere with transmissions on the higher frequency channels. Theoutput signal from filter 459 is the base band signal y₀(t). With acombination of low pass filter 459 and high pass filter 215 coupled tosummer 213, cross-channel interference between the base band channel,channel 301-0, and higher frequency channels 301-1 through 301-K can beminimized or eliminated.

FIG. 4B shows a block diagram of an embodiment of up-convertingmodulator 212-k, an arbitrary one of up-conversion modulators 212-1through 212-K of transmitter 270-t (FIG. 2C). Up-converting modulator212-k receives n_(k) bits per baud interval, 1/B_(k), for transmissioninto sub-channel 301-k. The parameter B_(k) denotes the baud rate, orsymbol rate, of the transmission. The n_(k) bits are received inscrambler 401. Scrambler 401 scrambles the n_(k) bits and outputs ascrambled signal of n_(k) bits, which whitens the data.

The output signal of n_(k) parallel bits from scrambler 401 is theninput to encoder 402. Although any encoding scheme can be utilized,encoder 402 can be a trellis encoder for the purpose of providing errorcorrection capabilities. Trellis coding allows for redundancy in datatransmission without increase of baud rate, or channel bandwidth.Trellis coding is further discussed in, for example, BERNARD SKLAR,DIGITAL COMMUNICATIONS, FUNDAMENTALS AND APPLICATIONS (Prentice-Hall,Inc., 1988), G. Ungerboeck., “Trellis Coding Modulation with RedundantSignal Sets, Part I. Introduction,” IEEE Communications Magazine, vol.25, no. 2, February 1987, pp. 5-11, and G. Ungerboeck., “Trellis CodingModulation with Redundant Signal Sets, Part II. State of the Art,” IEEECommunications Magazine, vol. 25, no. 2, February 1987, pp. 12-21. Otherencoding schemes include block coding schemes such as Reed-Solomonencoders, and BCH encoders, see, e.g., G. C. CLARK, JR., AND J. B.CAIN., ERROR CORRECTION CODING FOR DIGITAL COMMUNICATIONS (Plenum Press,New York, 1981), however they result in an increase of channel bandwidthusage. Typically, the signal output from encoder 402 includes more bitsthan n_(k), n_(k)+1e. In some embodiments, encoder 402 can be a trellisencoder which adds one additional bit, in other words encoder 402 can bea rate n_(k)/n_(k)+1 encoder, see, e.g., G. Ungerboeck., “Trellis CodingModulation with Redundant Signal Sets, Part I. Introduction,” IEEECommunications Magazine, vol. 25, no. 2, February 1987, pp. 5-11, and G.Ungerboeck., “Trellis Coding Modulation with Redundant Signal Sets, PartII. State of the Art,” IEEE Communications Magazine, vol. 25, no. 2,February 1987, pp. 12-21. In some embodiments, additional bits can beadded to insure a minimum rate of transitions so that timing recoverycan be efficiently accomplished at receiver 220-p. Typically, theencoder is referred to as an n_(k)/n_(k)+1e encoder.

The output signal from encoder 402 is input to symbol mapper 403. Symbolmapper 403 can include any symbol mapping scheme for mapping theparallel bit signal from encoder 402 onto symbol values fortransmission. In some embodiments, symbol mapper 403 is a QAM mapperwhich maps the (n_(k)+1e) bits from encoder 402 onto a symbol set withat least 2^((n) ^(k) ^(+1e)) symbols. A trellis encoder for encoder 402in conjunction with a QAM mapper for symbol mapper 403 can provide atrellis encoded QAM modulation for sub-channel 301-k.

The encoded output bits from encoder 402 are input to mapper 403. In anexample where n_(k)=6 and 1e=1, 7 bits from encoder 402 are input tomapper 403. If encoder 402 is the 16 state, rate 2/3 encoder discussedabove, the 3 bit output of encoder 402 can be the 3 most-significantbits (MSBs) and the 4 uncoded bits can be the least-significant bits(LSBs).

In some embodiments, a 16 symbol QAM scheme can be utilized. In thoseembodiments, 4 bits with no encoding (or 3 bits in an 3/4 encodingscheme) can be directly mapped onto 16 QAM symbols. In some embodiments,4 bits can be encoded (with a 4/5 encoding scheme) into a 32 QAM symbolset. In general, any size symbol set can be utilized.

The output signal from symbol mapper 403 can be a complex signalrepresented by in-phase signal I_(k)(ν) and a quadrature signalQ_(k)(ν), where ν is an integer that represents the νth clock cycle ofthe clock signal CK1, whose frequency equals the baud rate B_(k). Eachof signals I_(k)(ν) and Q_(k)(ν) are digital signals representing thevalues of the symbols they represent. In some embodiments, a QAM mapperonto a constellation with 128 symbols can be utilized. Otherconstellations and mappings are well known to those skilled in the art,see, e.g., BERNARD SKLAR, DIGITAL COMMUNICATIONS, FUNDAMENTALS ANDAPPLICATIONS (Prentice-Hall, Inc., 1988) and E. A. LEE AND D. G.MESSERSCHMITT, DIGITAL COMMUNICATIONS (Kluwer Academic Publishers,1988). The number of distinct combinations of I_(k)(ν) and Q_(k)(ν),then, represents the number of symbols in the symbol set of the QAMmapping and their values represent the constellation of the QAM mapping.Further examples of QAM symbol sets include 16 QAM symbol sets (16-QAM)and 4/5 encoded 32-QAM symbol sets (4/5 encoded 32 QAM).

The signals from symbol mapper 403, I_(k)(ν) and Q_(k)(ν), are input todigital-to-analog converters (DACs) 406 and 407, respectively. DACs 406and 407 operate at the same clock rate as symbol mapper 403. In someembodiments, therefore, DACs 406 and 407 are clocked at the symbol rate,which is the transmission clock frequency B_(k). The analog outputsignals from DACs 406 and 407, represented by I_(k)(t) and Q_(k)(t),respectively, can be input to low-pass filters 408 and 409,respectively. Low pass filters 408 and 409 are analog filters that passthe symbols represented by I_(k)(t) and Q_(k)(t) in the base band whilerejecting the multiple frequency range reflections of the base bandsignal.

The output signals from low-pass filters 408 and 409, designated I_(k)^(LPF)(t) and Q_(k) ^(LPF)(t), respectively, are then up-converted to acenter frequency f_(k) to generate the output signal of y_(k)(t), thekth channel signal. The output signal from low-pass filter 408, I_(k)^(LPF)(t), is multiplied by cos(2πf_(k)t) in multiplier 410. The outputsignal from low-pass filter 409, Q_(k) ^(LPF)(t), is multiplied bysin(2πf_(k)t) in multiplier 411. The signal sin(2πf_(k)t) can begenerated by PLL 414 based on the reference clock signal and the signalcos(2πf_(k)t) can be generated by a π/2 phase shifter 413.

However, because mixers 410 and 411 are typically not ideal mixers andthe sine wave input to mixer 410, and the resulting cosine wave input tomixer 411, often varies from a sine wave; signals having harmonics ofthe frequency f_(k) are also produced. Often, the harmonic signals inputto mixers 410 and 411 may more closely resemble square-wave signals thansine wave signals. Even if the “sine wave input” is a true sine wave,the most commonly utilized mixers, such as Gilbert Cells, may act as aband-limited switch, resulting in a harmonic signal with alternatingpositive and negative voltages with frequency the same as the “sine waveinput” signal. Therefore, the output signals from filters 408 and 409are still multiplied by signals that more closely resemble square wavesthan sine waves. As a result, signals having frequency 2f_(k), 3f_(k), .. . are also produced, as well as signals in the base band (0f_(k)).Although the amplitude of these signals may be attenuated with higherharmonics, they are non-negligible in the output signal. Additionally,even harmonics (i.e., 0f_(k), 2f_(k), 4f_(k) . . . ) are absent if theduty cycle of the harmonic sine wave input to mixers is 50%. Otherwise,some component of all of the harmonics will be present.

The output signals from multipliers 410 and 411 are summed in summer 412to form

$\begin{matrix}{{y_{k}(t)} = {{\xi_{k}^{0}{I_{k}^{LPF}(t)}} - {\zeta_{k}^{0}{Q_{k}^{LPF}(t)}} + {\sum\limits_{n > 0}{\left( {{\xi_{k}^{n}I_{k}^{LPF}{\cos\left( {2\pi\;{nf}_{k}t} \right)}} - {\zeta_{k}^{n}Q_{k}^{LPF}{\sin\left( {2\pi\;{nf}_{k}t} \right)}}} \right){\left( {k \geq 1} \right).}}}}} & (1)\end{matrix}$where ξ_(k) ^(n) and ζ_(k) ^(n) is the contribution of the nth harmonicto y_(k)(t). If the duty cycle of the harmonic input signals to mixers410 and 411 is near 50%, the even harmonics are low and the oddharmonics are approximately given by ξ_(k) ^(n)=1/n and ζ_(k) ^(n)=1/n .

The overall output of transmitter 210-p (FIG. 2B), the output fromsummer 216, is then given by

$\begin{matrix}{{z(t)} = {\sum\limits_{k = 0}^{K}{{y_{k}(t)}.}}} & (2)\end{matrix}$

In an example where the frequencies f₁ through f_(K) are given byfrequencies f₀ through (Kf₀), respectively, then, the overall outputsignal z(t) from transmitter 210-p is given by:

$\begin{matrix}\begin{matrix}{{z(t)} = {{y_{0}(t)} + {\sum\limits_{k = 1}^{K}\left( {{\xi_{k}^{0}{I_{k}^{LPF}(t)}} - {\zeta_{k}^{0}Q_{k}^{LPF}}} \right)} + {\xi_{1}^{1}{I_{1}^{LPF}(t)}\cos\;\omega_{0}t} -}} \\{{\zeta_{1}^{1}{Q_{1}^{LPF}(t)}\sin\;\omega_{0}t} + {\left( {{\xi_{1}^{2}{I_{1}^{LPF}(t)}} + {\xi_{2}^{1}{I_{2}^{LPF}(t)}}} \right)\cos\; 2\omega_{0}t} -} \\{{\left( {{\zeta_{1}^{2}{Q_{1}^{LPF}(t)}} + {\zeta_{2}^{1}{Q_{2}^{LPF}(t)}}} \right)\sin\; 2\omega_{0}t} +} \\{{\left( {{\xi_{1}^{3}{I_{1}^{LPF}(t)}} + {\zeta_{3}^{1}{I_{3}^{LPF}(t)}}} \right)\cos\; 3\omega_{0}t} -} \\{{\left( {{\zeta_{1}^{3}{Q_{1}^{LPF}(t)}} + {\zeta_{3}^{1}{Q_{3}^{LPF}(t)}}} \right)\sin\; 3\omega_{0}t} +} \\{{\left( {{\xi_{1}^{4}{I_{1}^{LPF}(t)}} + {\xi_{2}^{2}{I_{2}^{LPF}(t)}} + {\xi_{4}^{1}{I_{4}^{LPF}(t)}}} \right)\cos\; 4\omega_{0}t} -} \\{{\left( {{\zeta_{1}^{4}{Q_{1}^{LPF}(t)}} + {\zeta_{2}^{2}{Q_{2}^{LPF}(t)}} + {\zeta_{4}^{1}{Q_{4}^{LPF}(t)}}} \right)\cos\; 4\omega_{0}t}\; + \ldots} \\{= {{y_{0}(t)} + {\sum\limits_{k = 1}^{K}\left( {{\xi_{k}^{0}{I_{k}^{LPF}(t)}} - {\zeta_{k}^{0}{Q_{k}^{LPF}(t)}}} \right)} +}} \\{\sum\limits_{M = 1}^{\infty}{\sum\limits_{{\forall k},{{m \in {k^{*}n}} = M}}\left( {{\xi_{k}^{n}{I_{k}^{LPF}(t)}\cos\; M\;\omega_{0}t} - {\zeta_{k}^{n}{Q_{k}^{LPF}(t)}\sin\; M\;\omega_{0}t}} \right)}}\end{matrix} & (3)\end{matrix}$where ω₀ is 2πf₀ and where I_(k) ^(LPF)(t) and Q_(k) ^(LPF)(t) are 0 forall k>K.

As shown in Equation 3, the signal on channel one is replicated into allof the higher K channels, the base-band, and into harmonic frequenciesbeyond the base-band and the K channels. Filter 215 can remove thecontribution to the base-band channel from up-converting modulators212-1 through 212-K. The signal on channel two, for example, is alsotransmitted on channels 4, 6, 8, . . . , and the base-band. The signalon channel 3 is transmitted on channels 6, 9, 12, . . . and thebase-band. In general, the signal on channel k will be mixed intochannels 2 k, 3 k, . . . and the base-band. Further, the attenuation ofthe signals with higher harmonics in some systems can be such that thesignal from channel k is non negligible for a large number of harmonics,potentially up to the bandwidth of the process, which can be 30-40 GHz.

In some embodiments of the invention, a high pass filter 215 (see FIG.2C) receives the signal from summer 213. High pass filter 215 can, forexample, be a first-order high-pass filter with 3 dB attenuation atf₁/2. Filter 215 removes the DC harmonics, i.e. the base-bandtransmissions, from the transmitter. In embodiments with a separatebase-band transmission, then, cross-channel coupling into the base-bandis minimized or eliminated. Further, removing the base-band harmonicsfrom the transmitted signals simplifies cross-channel cancellation atreceiver 220-p. In embodiments where high pass filter 215 exists, mostof the base-band contribution from each of up-converting modulators212-1 through 212-K,

${\sum\limits_{k = 1}^{K}\left( {{\xi_{k}^{0}{I_{k}^{LPF}(t)}} - {\zeta_{k}^{0}{Q_{k}^{LPF}(t)}}} \right)},$is filtered out and becomes close to 0. The output signal fromtransmitter 210-p then becomes

$\begin{matrix}{{z^{\prime}(t)} = {{y_{0}(t)} + {\sum\limits_{M = 1}^{\infty}{\sum\limits_{{\forall k},{{n \in {k^{*}n}} = M}}{\left( {{\xi_{k}^{n}{I_{k}^{LPF}(t)}\cos\; M\;\omega_{0}t} - {\zeta_{k}^{n}{Q_{k}^{LPF}(t)}\sin\; M\;\omega_{0}t}} \right).}}}}} & (4)\end{matrix}$

In many embodiments the frequencies f₁ through f_(K) are chosen asmultiplies of a single frequency f₀ which can fulfill equations 3 and/or4 and results in the harmonic mixing of channels as shown in equation 3and 4. In embodiments that do not utilize a set of frequencies which aremultiples of a single frequency, f₀, cross-channel interference isimmensely more difficult to cancel.

In some embodiments of the invention, DACs 406 and 407 of the embodimentof up-converting modulator 212-k shown in FIG. 4B may be moved toreceive the output of summer 412. Further, in some embodiments DACs 406and 407 can be replaced by a single DAC to receive the output of summer213. However, such DACs should have very high sampling rates. Oneadvantage of utilizing high-sampling rate DACs is that ideal mixingcould take place and the number of harmonics that need to be cancelledcan be greatly reduced or even eliminated.

In some embodiments, DACs 406 and 407 of each of up-convertingmodulators 212-1 through 212-K can each be 4 bit DACs. The abovedescribed trellis encoder 402, in this embodiment, provides anasymptotic coding gain of about 6 dB over uncoded 128-QAM modulationwith the same data rate, see, e.g., G. Ungerboeck., “Trellis CodingModulation with Redundant Signal Sets, Part I. Introduction,” IEEECommunications Magazine, vol. 25, no. 2, February 1987, pp. 5-11, and G.Ungerboeck., “Trellis Coding Modulation with Redundant Signal Sets, PartII. State of the Art,” IEEE Communications Magazine, vol. 25, no. 2,February 1987, pp. 12-21.

As an example, then, embodiments of transmitter 210-p capable of 10 Gbpstransmission can be formed. In that case, η=10, i.e., an overallthroughput of 10 Gbps from the transmitter to the receiver. Someembodiments, for example, can have (K+1)=8 channels 301-0 through 301-7.Channels 301-1 through 301-7 can be 6/7 trellis encoded 128 QAM with thebaud rate on each channel B_(k) being 1.25 GHz/6 or about 208.333Msymbols/sec. Channel 301-0, the base-band channel, can be PAM-8 with noerror correction coding (i.e., uncoded PAM-8) with baud rate B₀ being416.667 Msymbols/sec. In other words, n_(k)=6; 1≦k≦7 and encoder 402 isa 6/7 rate trellis encoder. In this example, channels 301-1 through301-7 can be transmitted at frequencies 2f₀, 3f₀, 4f₀, 5f₀, 6f₀, 7f₀ and8f₀, respectively, where f₀ can be, for example, 1.5*B_(k) or 312.5 MHz.

In some embodiments of the invention, as shown in FIG. 2E, embodimentsof transceiver 255-p include two receivers (i.e., receivers 272-1 and272-2) and two transmitters (i.e., transmitters 270-1 and 270-2).Transceiver 255-p is coupled to a complementary transceiver 255-qthrough, for example, four pairs of category 5, 5E or 6 cabling, asshown in transmission medium 110 as media 110-1, 110-2, 110-3 and 110-4in FIG. 2E. As shown in FIG. 2E, transceiver 255-q also includes tworeceivers (i.e., receivers 272-1 and 272-2) and two transmitters (i.e.270-1 and 270-2) complementarily coupled to transceiver 255-p, i.e.transmitter 270-1 of transceiver 255-p is coupled through media 110-1 toreceiver 272-1 of transceiver 255-q; transmitter 270-2 of transceiver255-p is coupled through media 110-2 to receiver 272-2 of transceiver255-q; receiver 272-1 of transceiver 255-p is coupled through media110-3 to transmitter 270-1 of transceiver 255-q; and receiver 272-2 oftransceiver 255-p is coupled through media 110-4 to transmitter 270-2 oftransceiver 255-q.

In some embodiments as shown in FIG. 2E, each of transmitters 270-1 and270-2 of transceiver 255-p and transmitters 270-1 and 270-2 oftransceiver 255-q transmits utilizing four (4) channels with frequenciesf₀, 2f₀, 3f₀ and 4f₀, where f₀ is about 312.5 MHz (a factor of 1.5 timesthe baud rate) and a baud rate of 208.333 Msymbols/sec. The embodimentsof transceivers 255-p and 255-q can utilize QAM128 symbols with the 6/7trellis code. The resulting system can transmit a total of 10 Gbits/secin each direction between transceiver 255-p and 255-q.

FIG. 2F shows another embodiment of communicating complementarytransceivers 255-p and 255-q. In the embodiment shown in FIG. 2F,transceiver 255-p includes four (4) transmitters 270-1 through 270-4 andfour (4) receivers 272-1 through 272-4. Complementary transceiver 255-qalso includes four (4) transmitters 270-1 through 270-4 and four (4)receivers 272-1 through 272-4. Transmitters 270-1 through 270-4 oftransceiver 255-p are coupled to receivers 272-1 through 272-4 oftransceiver 255-q through four (4) media 110-1 through 110-4 oftransmission medium 110. Conversely, transmitters 270-1 through 270-4 oftransceiver 255-q are coupled to receivers 272-1 through 272-4 oftransceiver 255-p through the same four (4) media 110-1 through 110-4.Each of media 110-1 through 110-4 carries full-duplex data, i.e. dataflows both directions. Therefore, transmitter 270-1 of transceiver 255-pis coupled to receiver 272-1 of transceiver 255-q and receiver 272-1 oftransceiver 255-p is coupled to transmitter 270-1 of transceiver 255-qthrough media 110-1; transmitter 270-2 of transceiver 255-p is coupledto receiver 272-2 of transceiver 255-q and receiver 272-2 of transceiver255-p is coupled to transmitter 270-2 of transceiver 255-q through media110-2; transmitter 270-3 of transceiver 255-p is coupled to receiver272-3 of transceiver 255-q and receiver 272-3 of transceiver 255-p iscoupled to transmitter 270-3 of transceiver 255-q through media 110-3;transmitter 270-4 of transceiver 255-p is coupled to receiver 272-4 oftransceiver 255-q and receiver 272-4 of transceiver 255-p is coupled totransmitter 270-4 of transceiver 255-q through media 110-4.

Media 110-1 through 110-4 can be, for example, category 5, 5E or 6copper pairs. Each of transmitters 270-1 through 270-4 of transceiver255-p and transmitters 270-1 through 270-4 of transceiver 255-q includestwo channels with a baud rate of 208.333 Msymbols/sec each. Again, f₀can be 312.5 MHz, or 1.5 times the baud rate. Again, a QAM128 symbol setmay be utilized with 6/7 trellis encoding. The resulting totaltransmission rate is 10 Gbits/sec in each direction.

In some embodiments of the configuration shown in FIG. 2F, transmitters270-1 through 270-4 of transceiver 255-p and the complementary receivers272-1 through 272-4 of transceiver 255-q operate with channels atdifferent frequencies than transmitters 270-1 through 270-4 oftransceiver 255-q and the complementary receivers 272-1 through 272-4 oftransceiver 255-p. For example, transmitters 270-1 through 270-4 oftransceiver 255-p may each transmit on channels with carrier frequenciesf₀ and 3f₀ while transmitters 270-1 through 270-4 of transceiver 255-qtransmit on channels with carrier frequencies 2f₀ and 4f₀. However, insome embodiments of the invention, transmitters 270-1 and 270-2 oftransceiver 255-p and transmitters 270-1 and 270-2 of transceiver 255-qtransmit using the same set of frequencies, f₀ and 2f₀.

In another example embodiment, 10 Gbps (η=10) can utilize (K+1)=2channels 301-0 and 301-1. Channel 301-1 can be, for example, 16 QAM withno error correction coding (i.e., uncoded 16-QAM) with baud rate B₁ of1.25 GHz and channel 301-0 can be, for example, 16-PAM with no errorcorrection coding (i.e., uncoded 16-PAM) with baud rate B₀ at 1.25 GHz.The baud rate for both the PAM channel and the QAM channel is then 1.25Gsps. The throughput is 5 Gbps each for a total transmission rate of 10Gbps. With an excess bandwidth of the channels of about 50%, the centerfrequency of the QAM channel can be f₁≧(1.5)*1.25 GHz or above about 1.8GHz.

In another example embodiment, 10 Gbps can utilize (K+1)=2 channels301-0 and 301-1 as in the previous example with channel 301-1 being a4/5 trellis encoded 32 QAM with a baud rate B₁ of 1.25 GHz with channel301-0 being uncoded 16-PAM with baud rate B₀ of 1.25 GHz. Again, thecenter frequency of channel 301-1 can be f₁≧(1.5)*1.25 GHz or aboveabout 1.8 GHz.

In yet another example, (K+1)=6 channels, channels 301-0 through 301-5,can be utilized. Channels 301-1 through 301-5 can be 6/7 trellis encoded128-QAM with baud rate B_(k) of 1.25 GHz/6 or, 208 MHz. Channel 301-0,the base-band channel, can be 3/4 encoded 16 PAM or uncoded 8-PAM withbaud rate B₀=1.25 GHz. The center frequencies of channels 301-1 through301-5 can be 4f₀, 5f₀, 6f₀, 7f₀, and 8f₀, respectively, with f₀ beingabout 312.5 MHz.

Although several different embodiments are specifically described here,one skilled in the art will recognize that many other configurations oftransceivers according to the present invention are possible. One canarrange the number of transmitters and complementary receivers and thenumber of channels per transmitter in any way, depending on the desiredbaud rate and transceiver complexity. It is recognized that a givendesired baud rate can be achieved by providing transceivers with a smallnumber of transmitters transmitting on a large number of channels or alarger number of transmitters on a smaller number of channels. Further,data can be transmitted in half-duplex or full-duplex mode over thetransmission medium. Again, the filtering demands will be different forhalf-duplex or full-duplex transmission. For example, Echo filteringshould be included in full-duplex transmission, but may not be useful inhalf-duplex transmission.

As shown in FIGS. 2B and 2C, signals from base-band modulator 217 andeach of up-converting modulators 212-1 through 212-K of each oftransmitters 270-1 through 270-T are input to NEXT/Echo filter 250. Inthe embodiments shown in FIGS. 4A and 4B, the output signals from symbolmappers 456 and 403, respectively, are input to NEXT/Echo filter 250.Other embodiments of the invention may utilize other signals frombase-band modulator 217 and up-converting modulators 212-1 through 212-Kin order to correct for near-end cross talk and echo.

FIG. 5 shows a block diagram of an embodiment of receiver 272-r (seeFIGS. 2B and 2D). Receiver 272-r is included in receiver portion 220-pof transceiver 255-p and therefore is adjacent to transmitters 270-1through 270-T of transmitter portion 210-p of transceiver 255-p, asshown in FIG. 2B, which results in near-end cross talk interference(NEXT). Furthermore, receiver 272-r receives data from a complementarytransmitter over transmission medium 110. The complementary transmitteris adjacent to other transmitters, which results in far-end cross talkinterference (FEXT). Further, if transmission medium 110 is full duplex,then one of transmitters 270-1 through 270-T transmits over the samemedium on which receiver 272-r receives, which results in Echointerference. Additional interference results from the cross-channelinterference discussed above and the inter-symbol interference due totransmission medium 110 itself.

As shown in FIG. 2D, receiver 272-r of receiver portion 220-p includesdemodulators 221-1 through 221-K and possibly base-band demodulator 223to form a (K+1)-channel receiver. Some embodiments do not includebase-band demodulator 223. The output signals from receiver input buffer224, Z(t), are received in each of down-converting demodulators 222-1through 222-K and base-band demodulator 223 of receiver 272-r. Thesignal Z(t), then, is the transmitted signal z(t) from the complementarytransmitter after transmission through transmission medium 110. As shownin FIG. 3, the attenuation of signals at each of the K carrierfrequencies utilized by transmitter 272-r after transmission throughmedium 110 can be different. Additionally, the signal Z(t) suffers frominter-symbol interference caused by the dispersive effects of medium110.

The dispersive effects cause the signals received within a particulartiming cycle to be mixed with those signals at that carrier frequencyreceived at previous and future timing cycles. Therefore, in addition tocross-channel interference effects caused by the harmonic generation inmixers of the complementary transmitter (which is a transmitter of oneof transmitter portions 210-1 through 210-P), but also the signals foreach channel are temporally mixed through dispersion effects in medium110. Also, the signal received at input buffer 224 includes near-endcross talk and echo interference from transmitters 270-1 through 270-Tof transmitter portion 210-p of transceiver 255-p (see FIG. 2B).Further, FEXT interference results from the interference from the othertransmitters at the far end (i.e., the complementary transceiver)communicating with receiver 272-1 thorough 272-R. Therefore, the inputsignal Z(t) at input buffer 224 includes the signal transmitted by thecomplementary transmitter a contribution from the output signalstransmitted by transmitters 270-1 through 270-T and a signal transmittedby other transmitters at the complementary transceiver.

FIG. 5 shows embodiments of base-band demodulator 223 anddown-converting demodulators 222-1 through 222-K of transmitter 272-r.Signal Z(t) is received into each of base-band demodulator 223 anddown-converting demodulators 222-1 through 222-K. As shown in FIG. 5,down-converting demodulator 222-k, an arbitrary one of down-convertingdemodulators 222-1 through 222-K, for example, receives the signal Z(t)into down converter 560-k which down converts the channel transmitted atfrequency f_(k) back into the base-band and recovers in-phase andquadrature components Z_(k) ^(I)(t) and Z_(k) ^(Q)(t), respectively.

Down converter 560-k down converts the signal from Z(t) by a frequency{circumflex over (ƒ)}_(k), where {circumflex over (ƒ)}_(k) can be thelocally generated estimate of the carrier center frequency f_(k) fromthe complementary transmitter. The clock signals within component 201-p,an arbitrary one of components 201-1 through 201-P, which are generatedbased on the reference signal from PLL 230 as shown in FIG. 2A, willhave the same frequencies. However, the frequencies between differingones of components 201-1 through 201-P can be slightly different.Therefore, {f_(k)}denotes the set of frequencies at the complementarytransmitter and {{circumflex over (ƒ)}_(k)} denotes the set offrequencies at receiver 272-r. In some embodiments, the frequenciesutilized for each of the channels for transceiver 255-p as a whole arefixed.

As shown in FIG. 5, PLL 523 generates the clock signals for each ofdown-converting demodulators 222-1 through 222-K and base-banddemodulator 223 and, in particular, generates the sin(2π{circumflex over(ƒ)}_(k)t) signal for down-converting demodulator 222-k. Thecos(2π{circumflex over (ƒ)}_(k)t) signal can be generated by the π/2phase shifter in PLL 523. PLL 523 generates the sampling clock signalutilized in analog to digital converters (ADCs) 506-k and 507-k as wellas other timing signals utilized in down-converting demodulators 222-1through 222-K and base-band demodulator 223. PLL 523 also generates anRX CLK signal for output with the n_(k) bit output signal fromdown-converting demodulator 222-k of receiver 272-r.

PLL 523 can be a free-running clock for receiver 272-r based on areference clock signal. In some embodiments the complementarytransmitter of receiver 272-r of the receiver system 220-p, because theyare part of different ones of components 201-1 through 201-P, are atdifferent clock signals. This means that the PLLs for timing recoveryand carrier recovery correct both phase and frequency offsets betweenthe transmitter clock signals and receiver clock signals. Within one ofcomponents 201-1 through 201-P, each of the individual transmitters andreceivers can operate with the same PLL and therefore will operate withthe same clock signals. Components 201-i and 201-j, where i and j referto different ones of components 201-1 through 201-P, in general mayoperate at different clock signal frequencies.

In some embodiments, transceiver 255-p communicates with only onecomplementary transceiver 255-q, for example as shown in FIGS. 2E and2F. In such arrangements, transceiver 255-p can include a master clockthat controls the overall frequency of both transceiver 255-p and 255-q.In this arrangement, the clock of transceiver 255-p has a free-runningfrequency based on a reference clock. Transceiver 255-q is then slavedto the timing of transceiver 255-p. In other words, PLL 523 andfrequency shift 564 of transceiver 255-q are utilized to recover thefrequency of transceiver 255-p. Since the harmonic frequencies and baudrate frequencies are functions of each other, recovery of one frequencyfor one channel can lead to correct frequencies for all the mixers andthe baseband/baud rate of transceiver 255-q. Not only will the slave usethe recovered frequency to receive in its mixer and ADC and basebandcircuitry, it will also use this frequency in the transmit mixer and thebaseband transmit. This way, the master transceiver will automaticallyhave the correct frequency using its original clock. By having theentire system (baseband and mixers) run from one base frequency, thecomplexity of the filters such as Next/Echo filter 250, FEXT filter 251,or cross-channel interference filter 570 can be as described below.Embodiments of the invention where the entire system does not operatefrom one base frequency, the complexity of Next/Echo filter 250, FEXTfilter 250, and cross-channel interference filter 570 is much increased.

In more complex systems where transceiver 255-p may transmit tocomplementary receivers in more than one other transceiver and wheretransceiver 255-p may receive from complementary transmitters in morethan one other transceiver, a master/slave arrangement may also bearranged. The transmitted signals from each of transmitters 270-1through 270-T of transceiver 255-p may be utilized in each transceiverhaving a complementary receiver (i.e., a receiver that receives datafrom one of transmitters 270-1 through 270-T of transceiver 255-p) toset its internal frequency. The internal operating frequency fortransceivers that do not receive data from transceiver 255-p (i.e., onesthat do not include a complementary transceiver) can be slaved to thefrequencies transmitted by complementary transceivers that are slaved totransceiver 255-p. In that fashion, each of transceivers 255-1 through255-P can effectively be slaved to one transceiver 255-p.

Down converters 560-1 through 560-K also generate harmonics for verymuch the same reasons that harmonics are generated in up-convertingmodulators 212-1 through 212-K. Therefore, down converter 560-k willdown-convert into the base-band signals from signals having centerfrequencies 0, {circumflex over (ƒ)}_(k), 2{circumflex over (ƒ)}_(k),3{circumflex over (ƒ)}_(k), . . . . Additionally, the output signals Z₂^(I)(t) and Z₂ ^(Q)(t) include contributions from channels withfrequencies 0, 2{circumflex over (ƒ)}₀, 4{circumflex over (ƒ)}₀,6{circumflex over (ƒ)}₀ . . . and those channels with harmonics at thesefrequencies. Therefore, signals from channel k=3 need to be cancelledfrom signals transmitted on channel k=2. Each of the channels alsoinclude the cross-channel interference generated by the transmittermixers and the dispersive interference created by the channel. If thebase-band component of the harmonics is not filtered out by filter 215(FIG. 2C), then every channel could put a copy of its transmit signalonto the base-band and every channel will receive the base-band signalat the receive side.

The signals Z_(k) ^(I)(t) and Z_(k) ^(Q)(t) also will includeinterference from near-end cross talk and possibly Echo, i.e.interference from transmissions from transmitters 270-1 through 270-T intransmitter portion 210-p, and FEXT, i.e. interference fromtransmissions from transmitters transmitting to receivers 272-1 through272-R (except signals transmitted to receiver 272-r shown in FIG. 5).Additionally, all of the cross-channel effects in the transmission willmanifest in the interference of individual signals in down-convertingdemodulators 222-1 through 222-K and base-band demodulator 223.Therefore, in some embodiments the signals Z_(k) ^(I)(t) and Z_(k)^(Q)(t) output from down converter 560-k suffer the effects ofcross-channel interference resulting from harmonic generation in thetransmitter mixers of the complimentary transmitter transmitting toreceiver 272-r, the effects of cross-channel interference resulting fromharmonic generation in the receiver mixers of receiver 272-r, theeffects of near-end cross talk interference and echo, the effects of farend cross talk, and the effects of temporal, intersymbol interference,resulting from dispersion in the transport media. As an additionalcomplicating factor, in some embodiments the transmitter and receiverclocks of a complimentary communicating transmitter/receiver pair can bedifferent. Therefore, as an example, in embodiments where f₁ throughf_(K) of the transmitter correspond to frequencies f₀ through Kf₀,respectively, then {circumflex over (ƒ)}₁ through {circumflex over(ƒ)}_(K) of the receiver will correspond to frequencies (f₀+Δ) throughK(f₀+Δ), where Δ represents the frequency shift between PLL 523 ofreceiver 220-p and the PLL of the transmitter component. The transmittermixers then cause cross-channel interference by mixing the signalstransmitted at frequency f_(k) into 0f_(k), 2f_(k), 3f_(k) . . . (0,2kf₀, 3kf₀ . . . in one example). The receiver mixers causecross-channel interference by down-converting the signals received at0{circumflex over (ƒ)}_(k), {circumflex over (ƒ)}_(k), 2{circumflex over(ƒ)}_(k), 3{circumflex over (ƒ)}_(k). to the base-band. If thefrequencies {circumflex over (ƒ)}₀ is f₀+Δ, then the harmonics will bedown-converted to a base-band shifted in frequency by kΔ, 2kΔ, 3kΔ, . .. , respectively.

In some embodiments of the invention, receiver 220-p includes afrequency shift 564 which supplies a reference clock signal to PLL 523.The reference clock signal supplied to PLL 523 can be frequency shiftedso that Δ becomes 0. The frequency supplied to PLL 523 by frequencyshift 564 can be digitally created and the input parameters to frequencyshift 564 can be adaptively chosen to match the receiver frequency withthe transmitter frequency. In a master/slave environment, the frequencyis adjusted only if receiver 272-r is not part of the mastertransceiver.

The output signals from down-converter 560-k, Z_(k) ^(I)(t) and Z_(k)^(Q)(t), are input to analog filter 561-k. In some embodiments, analogfilter 561-k can provide offset correction to correct for any leakageonto signal Z(t) from the sine and cosine signals provided by PLL 523,plus any DC offset in filters 504-k and 505-k and ADCs 506-k and 507-k.The DC offset values can be adaptively chosen and, in some embodiments,after an initial start-up procedure, the DC offset values can be fixed.Analog filter 561-k can, in some embodiments, provide filtering forfiltering out signals in Z_(k) ^(I)(t) and Z_(k) ^(Q)(t) that are notassociated with signals at the base-band of down-converting demodulator222-k. Additionally, an amplification may be provided in filter 561-k.The gains of filter 561-k can be determined by an automatic gain control(AGC). The output signals from analog filter 561-k, then, can ber _(k) ^(I)(t)=LPF[Z(t)cos(2π{circumflex over (ƒ)}_(k) t)]g _(k) ^(1(I))r _(k) ^(Q)(t)=LPF[Z(t)sin(2π{circumflex over (ƒ)}_(k) t)]g _(k)^(1(Q)),  (5)where g_(k) ^(1(I)) and g_(k) ^(1(Q)) represents the gains of theamplification and Z_(k) ^(I)(t)=Z(t)cos(2π{circumflex over (ƒ)}_(k)t)and Z_(k) ^(Q)(t)=Z(t)sin(2π{circumflex over (ƒ)}_(k)t).

The signals output from analog filter 561-k, signals r_(k) ^(I)(t) andr_(k) ^(Q)(t), are input to analog-to-digital converters (ADC) 506-k and507-k, respectively, which forms digitized signals R_(k) ^(I)(ν) andR_(k) ^(Q)(ν) corresponding with the analog signals r_(k) ^(I)(t) andr_(k) ^(Q)(t), respectively. The integer index ν indicates the number ofclock cycles of the system clock, which is usually operating at thetransmission symbol rate. In some embodiments, ADCs 506-k and 507-koperate at a sampling rate that is the same as the transmission symbolrate, e.g. the QAM symbol rate. In some embodiments, ADCs 506-k and507-k can operate at higher rates, for example twice the QAM symbolrate. The timing clock signal SCLK, as well as the sine and cosinefunctions of Equation 5, is determined by PLL 523. In outputs with η=10,K=4, n_(k)=6 and two transmission media, as described above, ADCs 506-kand 507-k can operate at a rate of about 208.333 Msymbols/sec or, inembodiments with K=8 and two transmission media, about 104.167Msymbols/sec. In some embodiments, ADCs 506-k and 507-k can be 8-bitADCs. However, for 128 QAM operation, anything more than 7 bits can beutilized. In some embodiments, the gain of amplifiers in analog filters560-k can be set by automatic gain control circuit (AGC) 520-k based onthe digital output signals from ADCs 506-k and 507-k, R_(k) ^(I)(ν) andR_(k) ^(Q)(ν), respectively.

The output signals from ADCs 506-k and 507-k, R_(k) ^(I)(ν) and R_(k)^(Q)(ν), respectively, are input to a first digital filter 562-k. Insome embodiments of the invention, the in-phase and quadrature datapaths may suffer from small differences in phase, denoted θ_(k) ^(c),and small differences in gain. Therefore, in some embodiments a phaseand amplitude correction is included in digital filter 562-k. In orderto correct the phase and amplitude between the in-phase and quadraturedata paths, one of the values R_(k) ^(I)(ν) and R_(k) ^(Q)(ν) is assumedto be of the correct phase and amplitude. The opposite value is thencorrected. The phase error can be corrected by using the approximationfor small θ_(k) ^(c) where sin θ_(k) ^(c) is approximately θ_(k) ^(c),and cos θ_(k) ^(c) is approximately one. As an example, assume that thevalue for R_(k) ^(I)(ν) is correct, the value for R_(k) ^(Q)(ν) is thencorrected. This correction can be implemented by subtracting the valueθ_(k) ^(c)R_(k) ^(I)(ν) from R_(k) ^(Q)(ν), for example. The amplitudeof R_(k) ^(Q)(ν) can also be corrected by adding a small portion η_(k)^(c) of R_(k) ^(Q)(ν). The values η_(k) ^(c) and θ_(k) ^(c) can beadaptively determined in adaptive parameters block 517-k. Additionally,an arithmetic offset can be implemented by subtracting an offset valuefrom each of R_(k) ^(I)(ν) and R_(k) ^(Q)(ν). The offset values can alsobe adaptively chosen in adaptive parameter block 517-k.

Further, a phase rotation circuit can also be implemented in firstdigital filter 562-k. The phase rotation circuit rotates both thein-phase and quadrature signals by an angle {circumflex over (θ)}_(k) ¹.The angle {circumflex over (θ)}_(k) ¹ can be adaptively chosen. In someembodiments, the phase rotation circuit can be implemented before theangle and amplitude correction circuits.

Finally, a digital equalizer can also be implemented in digital filter562-k. Digital filter 562-k can be any combination of linear anddecision feed-back equalizers, the coefficients of which can beadaptively chosen.

In the embodiment of the invention shown in FIG. 5, the complex adaptiveequalizer included in digital filter 562-k can counter the intersymbolinterference caused by frequency dependent channel attenuation, and thereflections due to connectors and vias that exist in communicationsystem 200 (which can be a backplane communication system, aninter-cabinet communication system, or a chip-to-chip communicationsystem) and both transmit and receive low pass filters. It should benoted that because of the frequency division multiplexing of datasignals, as is accomplished in transmitter system 210-p and receiversystem 220-p, the amount of equalization needed in any one of channels301-0 through 301-K is minimal. In some embodiments, such as the16-channel, 6 bit per channel, 10 Gbps example, only about 1-2 dB oftransmission channel magnitude distortion needs to be equalized. In 8channel embodiments, 3-4 dB of distortion needs to be equalized. Inother words, the number of taps required in a transport function for theequalizer can be minimal (e.g., 1-4 complex taps) in some embodiments ofthe present invention, which can simplify receiver 220-p considerably.In some embodiments of the invention, the equalizer can have any numberof taps. In some embodiments, NEXT/Echo filter 250 may be implementedbefore cross-channel interference filter 570 and FEXT filter 251.

As shown in FIG. 5, cross-channel interference canceller 570 removes theeffects of cross-channel interference from signals E₁ ^(I)(ν) and E₁^(Q)(ν) through E_(K) ^(I)(ν) and E_(K) ^(Q)(ν) output from digitalfilters 562-1 through 562-K. Cross-channel interference can result, forexample, from harmonic generation in the transmitter and receivermixers, as has been previously discussed. In some embodiments of theinvention, cross-channel interference filter 570 may be placed beforethe equalizer of digital filter 562-k.

The output signals from digital filter 562-2, E_(k) ^(I)(ν) and E_(k)^(Q)(ν), for each of down-converting demodulators 222-1 through 222-Kare input to cross-channel interference filter 570. For convenience ofdiscussion, the input signals E_(k) ^(I)(ν) and E_(k) ^(Q)(ν) arecombined into a complex value E_(k)(ν)=E_(k) ^(I)(ν)+iE_(k) ^(Q)(ν)(where i is √{square root over (−1)}). A sum of contributions from eachof the channels (i.e., each of signals E₁(ν) through E_(k−1)(ν),E_(k+1)(ν) through E_(K)(ν) and, in some embodiments, E₀(ν)), aresubtracted from the value of E_(k)(ν) for each channel. The complexvalue F_(k)(ν) is F_(k) ^(I)(ν)+iF_(k) ^(Q)(ν), representing thein-phase and quadrature output signals, with F₀(ν) representing the realvalue of the base-band demodulator 223, are output from cross-channelinterference filter 570. The output signals F_(1,r) through F_(K,r) canbe determined by

$\begin{matrix}{{\begin{pmatrix}F_{1,r}^{I} \\F_{2,r}^{I} \\\vdots \\F_{k,r}^{I} \\\vdots \\F_{K,r}^{I}\end{pmatrix} = {{\begin{pmatrix}Z^{- N} & 0 & \ldots & 0 & \ldots & 0 \\0 & Z^{- N} & \ldots & 0 & \ldots & 0 \\\vdots & \vdots & ⋰ & \vdots & \ldots & \vdots \\0 & 0 & \ldots & Z^{- N} & \ldots & 0 \\\vdots & \vdots & \ldots & \vdots & ⋰ & \vdots \\0 & 0 & \ldots & 0 & \ldots & Z^{- N}\end{pmatrix}\begin{pmatrix}E_{1,r}^{I} \\E_{2,r}^{I} \\\vdots \\E_{k,r}^{I} \\\vdots \\E_{K,r}^{I}\end{pmatrix}} - {{RE}\left( {\begin{pmatrix}0 & Q_{2,1}^{I,r} & \ldots & Q_{k,1}^{I,r} & \ldots & Q_{K,1}^{I,r} \\Q_{1,2}^{I,r} & 0 & \ldots & Q_{k,2}^{I,r} & \ldots & Q_{K,2}^{I,r} \\\vdots & \vdots & ⋰ & \vdots & \vdots & \vdots \\Q_{1,k}^{I,r} & Q_{2,k}^{I,r} & \ldots & 0 & \ldots & Q_{K,k}^{I,r} \\\vdots & \vdots & \vdots & \vdots & ⋰ & \vdots \\Q_{1,K}^{I,r} & Q_{2,K}^{I,r} & \ldots & Q_{k,K}^{I,r} & \ldots & 0\end{pmatrix}\begin{pmatrix}E_{1,r} \\E_{2,r} \\\vdots \\E_{k,r} \\\vdots \\E_{K,r}\end{pmatrix}} \right)}}}{\begin{pmatrix}F_{1,r}^{Q} \\F_{2,r}^{Q} \\\vdots \\F_{k,r}^{Q} \\\vdots \\F_{K,r}^{Q}\end{pmatrix} = {{\begin{pmatrix}Z^{- N} & 0 & \ldots & 0 & \ldots & 0 \\0 & Z^{- N} & \ldots & 0 & \ldots & 0 \\\vdots & \vdots & ⋰ & \vdots & \ldots & \vdots \\0 & 0 & \ldots & Z^{- N} & \ldots & 0 \\\vdots & \vdots & \ldots & \vdots & ⋰ & \vdots \\0 & 0 & \ldots & 0 & \ldots & Z^{- N}\end{pmatrix}\begin{pmatrix}E_{1,r}^{Q} \\E_{2,r}^{Q} \\\vdots \\E_{k,r}^{Q} \\\vdots \\E_{K,r}^{Q}\end{pmatrix}} - {{IM}\left( {\begin{pmatrix}0 & Q_{2,1}^{Q,r} & \ldots & Q_{k,1}^{Q,r} & \ldots & Q_{K,1}^{Q,r} \\Q_{1,2}^{Q,r} & 0 & \ldots & Q_{k,2}^{Q,r} & \ldots & Q_{K,2}^{Q,r} \\\vdots & \vdots & ⋰ & \vdots & \vdots & \vdots \\Q_{1,k}^{Q,r} & Q_{2,k}^{Q,r} & \ldots & 0 & \ldots & Q_{K,k}^{Q,r} \\\vdots & \vdots & \vdots & \vdots & ⋰ & \vdots \\Q_{1,K}^{Q,r} & Q_{2,K}^{Q,r} & \ldots & Q_{k,K}^{Q,r} & \ldots & 0\end{pmatrix}\begin{pmatrix}E_{1,r} \\E_{2,r} \\\vdots \\E_{k,r} \\\vdots \\E_{K,r}\end{pmatrix}} \right)}}}} & (6)\end{matrix}$where Z⁻¹ represents a once cycle delay and the additional designation rindicates the receiver 272-r. For convenience, the time index ν has beenomitted. The subscript r has been added to index to receiver 272-r. Insome embodiments, cross channel interference into the baseband is alsoincluded, in which case the F_(0,r) and E_(0,r) and transfer functionsdescribing contributions in the base-band and caused by the base-bandchannel are also included in Equation 6, otherwise all of Q_(k,0) ^(r)and Q_(0,k) ^(r), where Q_(k,l) ^(r)=(Q_(k,l) ^(I,r),Q_(k,l)^(Q,r)),will be zero.

The transfer function Q_(k,l) ^(r) can have any number of taps and, ingeneral, can be given byQ _(k,l) ^(I,r)=σ_(k,l) ^(0,I,r)+σ_(k,l) ^(1,I,r) Z ⁻¹+σ_(k,l) ^(2,I,r)Z ⁻²+ . . . +σ_(k,l) ^(M,I,r) Z ^(−M);Q _(k,l) ^(Q,r)=σ_(k,l) ^(0,Q,r)+σ_(k,l) ^(1,Q,r) Z ⁻¹+σ_(k,l) ^(2,Q,r)Z ⁻²+ . . . +σ_(k,l) ^(M,Q,r) Z ^(−M).  (7)In general, each of the functions Q_(k,l) ^(r) can have a differentnumber of taps M. The delay time is determined by N and can also bedifferent for each channel. In some embodiments, the number of taps Mfor each function Q_(k,l) ^(r) can be the same. In some embodiments,delays can be added in order to match the timing between all of thechannels. In some embodiments, functions Q_(k,l) ^(r) can include anynumber of delays.

The coefficients σ_(k,l) ^(0,I,r) through σ_(k,l) ^(M,I,r) and σ_(k,l)^(0,Q,r) through σ_(k,l) ^(M,Q,r) can be adaptively chosen incross-channel adaptive parameter block 581 as shown in FIG. 5 in orderto optimize the performance of receiver system 220-p. In someembodiments, M is chosen to be 5. In some embodiments, transfer functionQ_(k,l) ^(r) may be just 2 complex numbers, when M=0. Further discussionof cross channel interference canceller 570 and the adaptively chosencoefficients can be found in U.S. application Ser. No. 10/310,255, filedon Dec. 4, 2002, which is incorporated by reference into thisapplication in its entirety.

Therefore, in cross channel interference canceller 570 the cross channelinterference is subtracted from the output signals from digital filters562-1 through 562-K. The output signals from cross-channel interferencefilter 570, F_(0,r) through F_(K) _(r) _(,r) are input to FEXT filter251. FEXT filter 251 corrects for the interference caused bytransmitters adjacent to the transmitter coupled to receiver 272-r. Inembodiments where the adjacent transmitters are coupled to receiversadjacent to receiver 272-r, i.e. others of receivers 272-1 through 272-Rof transceiver 255-p, then the FEXT correction can be given by

$\begin{matrix}{{\begin{pmatrix}H_{1,r}^{I} \\H_{2,r}^{I} \\\vdots \\H_{k,r}^{I} \\\vdots \\H_{K,r}^{I}\end{pmatrix} = {{\begin{pmatrix}Z^{- N} & 0 & \ldots & 0 & \ldots & 0 \\0 & Z^{- N} & \ldots & 0 & \ldots & 0 \\\vdots & \vdots & ⋰ & \vdots & \ldots & \vdots \\0 & 0 & \ldots & Z^{- N} & \ldots & 0 \\\vdots & \vdots & \ldots & \vdots & ⋰ & \vdots \\0 & 0 & \ldots & 0 & \ldots & Z^{- N}\end{pmatrix}\begin{pmatrix}F_{1,r}^{I} \\F_{2,r}^{I} \\\vdots \\F_{k,r}^{I} \\\vdots \\F_{K,r}^{I}\end{pmatrix}} - {\sum\limits_{i = 1}^{R}{{RE}\left( {\begin{pmatrix}{Fe}_{1,1}^{I,i,r} & {Fe}_{2,1}^{I,i,r} & \ldots & {Fe}_{k,1}^{I,i,r} & \ldots & {Fe}_{K_{i},1}^{I,i,r} \\{Fe}_{1,2}^{I,i,r} & {Fe}_{2,2}^{I,i,r} & \ldots & {Fe}_{k,2}^{I,i,r} & \ldots & {Fe}_{K_{i},2}^{I,i,r} \\\vdots & \vdots & ⋰ & \vdots & \vdots & \vdots \\{Fe}_{1,k}^{I,i,r} & {Fe}_{2,k}^{I,i,r} & \ldots & {Fe}_{k,k}^{I,i,r} & \ldots & {Fe}_{K_{i},k}^{I,i,r} \\\vdots & \vdots & \vdots & \vdots & ⋰ & \vdots \\{Fe}_{1,K_{r}}^{I,i,r} & {Fe}_{2,K_{r}}^{I,i,r} & \ldots & {Fe}_{k,K_{r}}^{I,i,r} & \ldots & {Fe}_{K_{i},K_{r}}^{I,i,r}\end{pmatrix}\begin{pmatrix}F_{1,i} \\F_{2,i} \\\vdots \\F_{k,i} \\\vdots \\F_{K_{i},i}\end{pmatrix}} \right)}}}}{\begin{pmatrix}H_{1,r}^{Q} \\H_{2,r}^{Q} \\\vdots \\H_{k,r}^{Q} \\\vdots \\H_{K,r}^{Q}\end{pmatrix} = {{\begin{pmatrix}Z^{- N} & 0 & \ldots & 0 & \ldots & 0 \\0 & Z^{- N} & \ldots & 0 & \ldots & 0 \\\vdots & \vdots & ⋰ & \vdots & \ldots & \vdots \\0 & 0 & \ldots & Z^{- N} & \ldots & 0 \\\vdots & \vdots & \ldots & \vdots & ⋰ & \vdots \\0 & 0 & \ldots & 0 & \ldots & Z^{- N}\end{pmatrix}\begin{pmatrix}F_{1,r}^{Q} \\F_{2,r}^{Q} \\\vdots \\F_{k,r}^{Q} \\\vdots \\F_{K,r}^{Q}\end{pmatrix}} - {\sum\limits_{i = 1}^{R}{{{IM}\left( {\begin{pmatrix}{Fe}_{1,1}^{Q,i,r} & {Fe}_{2,1}^{Q,i,r} & \ldots & {Fe}_{k,1}^{Q,i,r} & \ldots & {Fe}_{K_{i},1}^{Q,i,r} \\{Fe}_{1,2}^{Q,i,r} & {Fe}_{2,2}^{Q,i,r} & \ldots & {Fe}_{k,2}^{Q,i,r} & \ldots & {Fe}_{K_{i},2}^{Q,i,r} \\\vdots & \vdots & ⋰ & \vdots & \vdots & \vdots \\{Fe}_{1,k}^{Q,i,r} & {Fe}_{2,k}^{Q,i,r} & \ldots & {Fe}_{k,k}^{Q,i,r} & \ldots & {Fe}_{K_{i},k}^{Q,i,r} \\\vdots & \vdots & \vdots & \vdots & ⋰ & \vdots \\{Fe}_{1,K_{r}}^{Q,i,r} & {Fe}_{2,K_{r}}^{Q,i,r} & \ldots & {Fe}_{k,K_{r}}^{Q,i,r} & \ldots & {Fe}_{K_{i},K_{r}}^{Q,i,r}\end{pmatrix}\begin{pmatrix}F_{1,i} \\F_{2,i} \\\vdots \\F_{k,i} \\\vdots \\F_{K_{i},i}\end{pmatrix}} \right)}.}}}}} & (8)\end{matrix}$Again, the transfer functions Fe_(k,l) ^(i,r), where Fe_(k,l)^(i,r)=(Fe_(k,l) ^(I,i,r),Fe_(k,l) ^(Q,i,r)), is the transfer functiondescribing the interference cancellation to the lth channel of the rthreceiver for signals from the kth channel of the ith receiver. FIG. 7shows, in block diagram form, a portion of FEXT filter 251. The portionof FEXT filter 251 shown in FIG. 7 is the interference correction forthe signal F_(k,r) for signals received at receivers 0 through R. Sinceno correction is made for signals received at receiver 272-r, thetransfer function Fe_(k,l) ^(r,r) is understood to be 0 and thereforethere are no blocks in FIG. 7 for FEXT interference correction fromreceiver 272-r. As shown in FIG. 7, and mathematically depicted inEquation 8, a correction is made to each of signals F_(0,1) throughF_(K,R) in transfer function blocks 703 and the results are summed insummer 701. The resulting total correction is subtracted from F_(k,r),which is delayed in delay block 704, in summer 702 to form the valueH_(k,r). This correction is calculated for each of signals F_(0,1)through F_(K,R) in transceiver 255-p.

In embodiments such as that shown in FIG. 7, corrections into and fromthe baseband is included in FEXT filter 251. If the baseband is includedin the filter calculations, equation 8 can be modified to includetransfer functions Fe_(k,0) ^(i,r) and Fe_(0,k) ^(i,r), input signalsF_(0,r), and output signals H_(0,r); otherwise, all of Fe_(k,0) ^(i,r)and Fe_(0,k) ^(i,r) are zero.

The transfer function Fe_(k,l) ^(i,r) can have any number of taps and,in general, can be given byFe _(k,l) ^(I,i,r)=β_(k,l) ^(0,I,i,r)+β_(k,l) ^(1,I,i,r) Z ⁻¹+β_(k,l)^(2,I,i,r) Z ⁻²+ . . . +β_(k,l) ^(W,I,i,r) Z ^(−W);Fe _(k,l) ^(Q,i,r)=β_(k,l) ^(0,Q,i,r)+β_(k,l) ^(1,Q,i,r) Z ⁻¹+β_(k,l)^(2,Q,i,r) Z ⁻²+ . . . +β_(k,l) ^(Q,W,i,r) Z ^(−W).  (9)The coefficients β_(k,l) ^(w,I or Q,i,r) can be adaptively chosenaccording toβ_(k,l) ^(w,x,I,i,r)(ν+1)=β_(k,j) ^(w,x,I,i,r)(ν)+α(e _(l,r) ^(I)(ν)F_(k,i) ^(I)(ν−w))β_(k,l) ^(w,y,I,i,r)(ν+1)=β_(k,l) ^(w,y,I,i,r)(ν)−α(e _(l,r) ^(I)(ν)F_(k,i) ^(Q)(ν−w))β_(k,l) ^(w,x,Q,i,r)(ν+1)β_(k,l) ^(w,x,Q,i,r)(ν)+α(e _(l,r) ^(Q)(ν)F_(k,i) ^(Q)(ν−w))β_(k,l) ^(w,y,Q,i,r)(ν+1)=β_(k,l) ^(w,y,Q,i,r)(ν)+α(e _(l,r) ^(Q)(ν)F_(k,i) ^(I)(ν−w)),  (10)where β_(k,l) ^(w,I,i,r)=β_(k,l) ^(w,x,I,i,r)+iβ_(k,l) ^(w,y,I,i,r) andβ_(k,l) ^(w,Q,i,r)=β_(k,l) ^(w,x,Q,i,r)+iβ_(k,l) ^(w,y,Q,i,r). Theparameter α controls the rate at which Equation 10 converges on valuesfor the coefficients β_(k,l) ^(w,I or Q,i,r) and the stability of thosevalues. In principle, α can be different for each of the adaptiveequations shown as Equation 10. The parameters e_(l,r) ^(I) and e_(l,r)^(Q) is the error between decided on symbols and input signals at slicer516-1 of receiver 272-r.

In some embodiments of the invention, cross channel interference filter570 and FEXT filter 251 can be combined into a single filter. As seen inEquations 6 and 8, Equation 8 can be easily modified to include thecross channel contributions described in Equation 6. The resultingcombinations would combine the adaptively chosen transfer functions intoa single transfer functions with adaptively chosen parameters, the newtransfer function being of the form of Fe_(k,l) ^(i,r) as described withEquation 8 above for FEXT filter 251 alone, except that with theadditional contributions from cross-channel interference as describedwith Equation 6 above the Fe_(k,l) ^(r,r) is not necessarily 0, onlyFe_(k,k) ^(r,r) is still understood to be 0. The adaptively chosenparameters would include contributions for corrections for cross-channelinterference and FEXT interference, but the resulting implementationwould likely be smaller than with separate cross-channel interferencefilter 570 and FEXT filter 251 as is shown in FIG. 5. The embodimentshown in FIG. 5 separates the two filters only in order to simplifydiscussions of the two different contributions to the signalinterference corrected by filters according to the present invention inreceiver 272-r.

As shown in FIG. 5, the output signals from FEXT interference filter ineach channel, H₀ and H₁ through H_(K) are input to NEXT/Echo filters250-0 through 250-K, respectively. For convenience in notation, thedesignation indicating receiver and the time index ν have been droppedand it is understood that receiver 272-r and time period ν areindicated. NEXT/Echo filters 250-0 through 250-K are further discussedbelow with reference to FIG. 6. NEXT/Echo filters 250-0 through 250-Kreceive signals from transmitters 270-1 through 270-T of transmitterportion 210-p of transceiver 255-p and removes the contribution ofsignals from transmitters 270-1 through 270-T from the values H₀ throughH_(K) in each of receivers 272-1 through 272-R that are due to near-endcross talk between transmitters 270-1 through 270-T and receivers 272-1through 272-R.

FIG. 6 shows a block diagram of an embodiment of NEXT/Echo filter250-k,r, which is a part of NEXT filter 250 shown in FIG. 2B. Inparticular, NEXT/Echo filter 250-k,r is an arbitrary one of NEXT/Echofilters 250-0 through 250-K on receiver 272-r, as shown in FIG. 5, eachof which is a part of NEXT/Echo filter 250 shown in FIG. 2B. NEXT/Echofilter 250-k,r receives signals from each of transmitters 270-1 through270-T of transmitter portion 210-p of transceiver 255-p. As shown inFIG. 6, signals S_(0,1) through S_(K) _(T) _(,T) are received intotransfer function blocks 601-(0,1) through 601-(K_(T),T), respectively,where K_(t) indicates the number of channels in transmitter 270-t.Signal S_(0,1) through S_(0,T) are base-band transmitted signals frombase-band modulators 217 from each of transmitters 270-1 through 270-T.As shown in FIG. 4A, signal S_(0,t) may be the output signal from symbolmapper 456 of transmitter 270-t.

Signals S_(1,t) through S_(K) _(t) _(,t) can be the complex outputsignals from up-converting modulators 212-1 through 212-K_(t) oftransmitter 270-t, respectively. The signal S_(k,t), for example, can bethe complex output signal from symbol mapper 403 of up-convertingmodulator 212-k (see FIG. 4B) of transmitter 270-t (see FIG. 2A).Therefore, S_(k,t)=I_(k) ^(t)+iQ_(k) ^(t), where k=1 to K_(t) and tindicates transmitter 270-t of transmitter portion 210-p of transceiver255-p.

Each of transfer functions FN_(0,k) ^(1,r) through FN_(K) _(T) _(,k)^(T,r) in blocks 601-(0,1) through 601-(K_(T),T) of NEXT/Echo filter250-k determines the amount of signal S_(0,1) through S_(K) _(T) _(,T),respectively, that should be removed from the signal H_(k) in receiver272-r (see FIG. 5) to correct for near-end cross talk and Echo ifnecessary. Function FN_(k) _(t) _(,k) _(r) ^(t,r), where k_(t) indexesthe k_(t)th channel, transmitted by up-converting modulator 212-k _(t)in transmitter 270-t, and runs between 1 and K_(t); and k_(r) indexesthe k_(r)th channel received in down-converting demodulator 222-k _(r)in receiver 272-r and runs between 1 and K_(r), represents an arbitraryone of the transfer functions of function blocks 601-(0,1) through601-(K_(T),T). The function FN_(k) _(t) _(,k) _(r) ^(t,r), where FN_(k)_(t) _(,k) _(r) ^(t,r) represents the combination (FN_(k) _(t) _(,k)_(r) ^(I,t,r),FN_(k) _(t) _(,k) _(r) ^(Q,t,r)), can be expressed asFN _(k) _(t) _(,k) _(r) ^(I,t,r)=δ_(k) _(t) _(,k) _(r) ^(0,I,t,r)+δ_(k)_(t) _(,k) _(r) ^(1,I,t,r) Z ⁻¹+ . . . +δ_(k) _(t) _(k) _(r) ^(w,I,t,r)Z ^(−w)+ . . . +δ_(k) _(t) _(,k) _(r) ^(W,I,t,r) Z ^(−W)FN _(k) _(t) _(,k) _(r) ^(Q,t,r)=δ_(k) _(t) _(,k) _(r) ^(0,Q,t,r)+δ_(k)_(t) _(,k) _(r) ^(1,Q,t,r) Z ⁻¹+ . . . +δ_(k) _(t) _(,k) _(r) ^(w,Q,t,r)Z ^(−w)+ . . . +δ_(k) _(t) _(,k) _(r) ^(W,Q,t,r) Z ^(−W),  (11)where δ_(k) _(t) _(,k) _(r) ^(w,I,t,r)=δ_(k) _(t) _(,k) _(r)^(w,x,I,t,r)+δ_(k) _(t) _(,k) _(r) ^(w,y,I,t,r) and δ_(k) _(t) _(,k)_(r) ^(w,Q,t,r)=δ_(k) _(t) _(,k) _(r) ^(w,x,Q,t,r)+iδ_(k) _(t) _(,k)_(r) ^(w,y,Q,t,r) can be complex coefficients and Z⁻¹ represents a onecycle delay. The value M is an integer representing the number of tapsin each function and can be different for each transfer functionrepresented. For Echo cancellation, the number of delays may beincreased to accommodate travel time of signals between complementarytransceivers. Further, for NEXT/Echo filter 250-0, corresponding to theNEXT filter in base-band demodulator 223, all the output signals for allfunctions FN_(0,0) ^(l,r) through FN_(K) _(T) _(,0) ^(T,r) may have realoutputs.

Therefore, the output signals from NEXT/Echo interference filters 250-1through 250-K_(r) of receiver 252-r can be represented as

$\begin{matrix}{{\begin{bmatrix}{Ne}_{1,r}^{I} \\{Ne}_{2,r}^{I} \\\vdots \\{Ne}_{k_{r},r}^{I} \\\vdots \\{Ne}_{K_{r},r}^{I}\end{bmatrix} = {{\begin{pmatrix}Z^{- D_{r}} & 0 & \ldots & 0 & \ldots & 0 \\0 & Z^{- D_{r}} & \ldots & 0 & \ldots & 0 \\\vdots & \vdots & ⋰ & \vdots & \ldots & \vdots \\0 & 0 & \ldots & Z^{- D_{r}} & \ldots & 0 \\\vdots & \vdots & \ldots & \vdots & ⋰ & \vdots \\0 & 0 & \ldots & 0 & \ldots & Z^{- D_{r}}\end{pmatrix}\begin{bmatrix}H_{1,r}^{I} \\H_{2,r}^{I} \\\vdots \\H_{k_{r},r}^{I} \\\vdots \\H_{K_{r},r}^{I}\end{bmatrix}} - {{Re}\left\lbrack {\begin{pmatrix}{FN}_{1,1}^{1,r} & {FN}_{2,1}^{1,r} & \ldots & {FN}_{k_{t},1}^{t,r} & \ldots & {FN}_{K_{t},1}^{T,r} \\{FN}_{1,2}^{1,r} & {FN}_{2,2}^{1,r} & \ldots & {FN}_{k_{t},2}^{t,r} & \ldots & {FN}_{K_{T},2}^{T,r} \\\vdots & \vdots & ⋰ & \vdots & \vdots & \vdots \\{FN}_{1,k_{r}}^{1,r} & {FN}_{2,k_{r}}^{1,r} & \ldots & {FN}_{k_{t},k_{r}}^{t,r} & \ldots & {FN}_{K_{t},k_{r}}^{T,r} \\\vdots & \vdots & \vdots & \vdots & ⋰ & \vdots \\{FN}_{1,K_{r}}^{1,r} & {FN}_{2,K_{r}}^{1,r} & \ldots & {FN}_{k_{t},K_{r}}^{t,r} & \ldots & {FN}_{K_{t},K_{r}}^{T,r}\end{pmatrix}\begin{bmatrix}S_{1,1} \\S_{2,1} \\\vdots \\S_{k_{t},t} \\\vdots \\S_{K_{T},T}\end{bmatrix}} \right\rbrack}}},{{{and}\begin{bmatrix}{Ne}_{1,r}^{Q} \\{Ne}_{2,r}^{Q} \\\vdots \\{Ne}_{k_{r},r}^{Q} \\\vdots \\{Ne}_{K_{r},r}^{Q}\end{bmatrix}} = {{\begin{pmatrix}Z^{- D_{r}} & 0 & \ldots & 0 & \ldots & 0 \\0 & Z^{- D_{r}} & \ldots & 0 & \ldots & 0 \\\vdots & \vdots & ⋰ & \vdots & \ldots & \vdots \\0 & 0 & \ldots & Z^{- D_{r}} & \ldots & 0 \\\vdots & \vdots & \ldots & \vdots & ⋰ & \vdots \\0 & 0 & \ldots & 0 & \ldots & Z^{- D_{r}}\end{pmatrix}\begin{bmatrix}H_{1,r}^{Q} \\H_{2,r}^{Q} \\\vdots \\H_{k_{r},r}^{Q} \\\vdots \\H_{K_{r},r}^{Q}\end{bmatrix}} - {{{Im}\left\lbrack {\begin{pmatrix}{FN}_{1,1}^{1,r} & {FN}_{2,1}^{1,r} & \ldots & {FN}_{k_{t},1}^{t,r} & \ldots & {FN}_{K_{t},1}^{T,r} \\{FN}_{1,2}^{1,r} & {FN}_{2,2}^{1,r} & \ldots & {FN}_{k_{t},2}^{t,r} & \ldots & {FN}_{K_{t},2}^{T,r} \\\vdots & \vdots & ⋰ & \vdots & \vdots & \vdots \\{FN}_{1,k_{r}}^{1,r} & {FN}_{2,k_{r}}^{1,r} & \ldots & {FN}_{k_{t},k_{r}}^{t,r} & \ldots & {FN}_{K_{t},k_{r}}^{T,r} \\\vdots & \vdots & \vdots & \vdots & ⋰ & \vdots \\{FN}_{1,K_{r}}^{1,r} & {FN}_{2,K_{r}}^{1,r} & \ldots & {FN}_{k_{t},K_{r}}^{t,r} & \ldots & {FN}_{K_{t},K_{r}}^{T,r}\end{pmatrix}\begin{bmatrix}S_{1,1} \\S_{2,1} \\\vdots \\S_{k_{t},t} \\\vdots \\S_{K_{T},T}\end{bmatrix}} \right\rbrack}.}}}} & (12)\end{matrix}$

In some embodiments of the invention, the coefficients of Equation 12can be fixed. In some embodiments, the coefficients of Equation 12 canbe adaptively chosen. In some embodiments, the coefficients can beadaptively chosen as follows:δ_(k) _(t) _(,k) _(r) ^(w,x,I,t,r)(ν+1)=δ_(k) _(t) _(,k) _(r)^(w,x,I,t,r)(ν)+α(e _(k) _(r) _(,r) ^(I)(ν)S _(k) _(t) _(,t) ^(I)(ν−w))δ_(k) _(t) _(,k) _(r) ^(w,y,I,t,r)(ν+1)=δ_(k) _(t) _(,k) _(r)^(w,y,I,t,r)(ν)−α(e _(k) _(r) _(,r) ^(I)(ν)S _(k) _(t) _(,t) ^(Q)(ν−w))δ_(k) _(t) _(,k) _(r) ^(w,x,Q,t,r)(ν+1)=δ_(k) _(t) _(,k) _(r)^(w,x,Q,t,r)(ν)+α(e _(k) _(r) _(,r) ^(Q)(ν)S _(k) _(t) _(,t) ^(Q)(ν−w))δ_(k) _(t) _(,k) _(r) ^(w,x,Q,t,r)(ν+1)=δ_(k) _(t) _(,k) _(r)^(w,y,Q,t,r)(ν)+α(e _(k) _(r) _(,r) ^(Q)(ν)S _(k) _(t) _(,t)^(I)(ν−w))  (13)The coefficient α, which in some embodiments can be different for eachof the parameters described by Equation 14, determines how fast equation14 converges on stable parameters. Such parameters are typically chosento be on the order of 10⁻³ to 10⁻⁵.

As indicated in Equations 11 and 12, the complex output signals fromeach of function blocks 601-(0,1) through 601-(K_(T),T) are summed insummer 602. The resulting sum is subtracted from the correspondingoutput signal from cross channel interference filter for channel 301-k(i.e., base-band demodulator 223 if k=0 or down-converting demodulator222-k if k>0) in summer 603.

The embodiment shown physically in FIG. 6 includes the contributionsfrom base-band modulator 217 of transmitters 270-1 through 270-T.However, some embodiments do not include the base-band contributions.Equation 12, for example, does not include that contribution, althoughone skilled in the art can easily modify Equation 13 to include thebase-band contributions.

As shown in FIG. 5, the output signals from NEXT/Echo filter 250-k,Ne_(k) ^(I) and Ne_(k) ^(Q), are input to digital filter 563-k. Digitalfilter 563-k can include further amplification, further offset, andquadrature correction. The amplification can be determined by automaticgain control such that, for example, the error signals generated byslicer 516-k are minimized. The offset values can also be set to reduceor minimize the errors at slicer 516-k. Further, a quadrature correctioncan be applied to correct for the phase error between the in-phase andquadrature mixers at the transmitter. The output signals from digitalfilter 563-k, G_(k) ^(I)(ν) and G_(k) ^(Q)(ν), then, can be given byG _(k) ^(I)(ν)=g _(k) ^(2-I) Ne _(k) ^(I)(ν)−OFFSET₂ ^(I)G _(k) ^(Q)(ν)=g _(k) ^(2-Q) Ne _(k) ^(Q)(ν)−g _(k) ^(2-I) Ne _(k)^(I)(ν){circumflex over (θ)}_(k) ⁽²⁾−OFFSET₂ ^(Q),  (14)where g_(k) ^(2(I)) and g_(k) ^(2(Q)) can be adaptively chosen gainvalues, {circumflex over (θ)}_(k) ⁽²⁾(ν) can be the adaptively chosenquadrature correction, and OFFSET₂ ^(I) and OFFSET₂ ^(Q) can be theadaptively chosen offsets.

Adaptive parameters 517-k receives signals from various sections ofdown-converting demodulator 222-k, including errors and decided onsymbol values from slicer 516-k, and adjusts various parameters indown-converting demodulator 222-k to optimize performance. Someparameters that can be adaptively chosen include gains, quadraturecorrections, offsets, and equalizer coefficients. The parameters forcross-channel interference filter 570 can be chosen in cross-channeladaptive parameter block 581. In some embodiments, the parameters chosenby cross-channel adaptive parameter block 581 are chosen based on thedecided on symbols ({circumflex over (α)}_(k,r) ^(I) and {circumflexover (α)}_(k,r) ^(Q)) and errors (e_(k,r) ^(I) and e_(k,r) ^(Q))generated in slicers 516-1 through 516-K.

In some embodiments, frequency shift 564 generates a reference signalinput to PLL 523 such that the frequency of component 201-p withreceiver system 220-p, {circumflex over (ƒ)}₁ through {circumflex over(ƒ)}_(K), matches the frequency of the corresponding component 201-qwith transmitter system 210-q, f₁ through f_(K), where component 201-qis transmitting data to component 201-p. In embodiments where f₁ throughf_(K) correspond to frequencies f₀ through Kf₀, respectively, thenfrequency shift 564 shifts the frequency of a reference clock such thatthe frequency shift Δ is zero. The frequencies {circumflex over (ƒ)}₁through {circumflex over (ƒ)}_(K), then, are also frequencies f₀ throughKf₀. In some embodiments, frequency shift 564 can receive input from anyor all loop filters in adaptive parameter blocks 577 and 517-1 through517-K and adjusts the frequency shift such that {circumflex over(θ)}_(k) ⁽¹⁾ through {circumflex over (θ)}_(k) ^((K)) remain a constant,for example 0 or any other angle. In some embodiments, frequency shift564 receives the output signals from any or all loop filters of adaptiveparameter blocks 517-1 through 517-K and baseband adaptive parameterblock 577. Frequency shift 564 can also change the frequency of thebaseband receive clock to match that of the far end transceiver.

In Master/Slave environments, the Master can assume that its frequencyshift can be 0 (for either just the mixers or for both the mixers andbaseband clocks) and thus do no correction. While the slave will notonly adjust its receive mixer and receive baseband clock to match themaster with frequency shift 564, but it will also adjust its transmitclock frequencies (again for either just the mixers or for both themixers and baseband clocks). By adjusting the transmit and receive mixerfrequencies, the complexity of the filters utilized in transceiver 255-pcan be as described as opposed to more complicated systems. By adjustingthe baseband frequencies as well, the Echo/NEXT filter also are asdescribed in the disclosure as opposed to an immensely more complicatedsystem.

As shown in FIG. 5, the output signals from digital filter 563-k,equalized samples {G_(k) ^(I)(ν), G_(k) ^(Q)(ν)}, are input to trellisdecoder 514-k. Trellis decoding can be performed using the Viterbialgorithm, see, e.g., G. Ungerboeck., “Channel Coding withMultilevel/Phase Signals,” IEEE Transactions on Information Theory, vol.IT-28, January 1982, pp. 55-67, G. Ungerboeck., “Trellis CodingModulation with Redundant Signal Sets, Part I. Introduction,” IEEECommunications Magazine, vol. 25, no. 2, February 1987, pp. 5-11, G.Ungerboeck., “Trellis Coding Modulation with Redundant Signal Sets, PartII. State of the Art,” IEEE Communications Magazine, vol. 25, no. 2,February 1987, pp. 12-21, or G. C. CLARK, JR., AND J. B. CAIN, ERRORCORRECTION CODING FOR DIGITAL COMMUNICATIONS, PP. 253-264 (Plenum Press,New York, 1981). Additionally, trellis decoder 514 converts from the QAMsymbol set back to parallel bits. The output signal from trellis decoder514, which now contains n_(k) parallel bits, is input to descrambler515-k. Descrambler 515-k of receiver down-converting demodulator 222-koperates to reverse the scrambling operation of scrambler 401 ofup-converting modulator 212-k.

As is shown in FIG. 2D, the output signals from each of down-convertingdemodulators 222-1 through 222-K and base-band demodulator 223 arerecombined into an N_(r) ^(R)-bit parallel signal in bit parsing 221.Additionally, the RX clock signal is output from bit parsing 221. Thedesignation N_(r) ^(R) indicates that there may be a different number ofbits (with a different bit distribution between modulators) for each oftransmitters 272-1 through 272-R, with N_(r) ^(R) indicating the numberof bits received by transmitter 272-r.

Base-band demodulator 223 shown in FIG. 5 also receives the signal Z(t)from medium 110. Analog processing 571 in base-band demodulator 223receives signal Z(t). Analog processing 571, for example, can include alow-pass filter in order to separate the base-band signal from thosesignals transported with carrier frequencies, such as those transmittedby up-converting modulators 212-1 through 212-K. Processor 571 canfurther include some analog correction of the signals, includinganti-aliasing filters, base-line wander filters, or other filters.

The output signal from analog processing 571 is input to ADC 572, whereit is digitized. ADC 572 can have any number of bits of resolution. Atleast a four bit ADC, for example, can be utilized in a 16-PAM system.ADC 572 can be clocked from a clock signal generated by receiver 120-pin general, for example in PLL 523 as shown in FIG. 5. In someembodiments, adaptive parameter control 577 can generate a phase signalwhich can add a phase to the timing of ADC 572.

The output signal from ADC 572 can be input to a digital filter 573.Further filtering and shaping of the signal can occur in digital filter573. Filter 573 can include, for example, a digital base-line wanderfilter, a digital automatic gain control circuit, or any other filter.The output signal from digital filter 573 can be input to cross-channelinterference filter 570, which cancels any interference with thebase-band signal processed by base-band demodulator 223 from theremaining channels 301-1 through 301-K. The output signal fromcross-channel interference filter 570 for channel 301-0, F₀, is theninput to FEXT filter 251. The output signal from FEXT filter 251, H₀, isthen input to NEXT/Echo interference filter 250-0. NEXT/Echointerference filter 250-0 cancels the interference to channel 301-0 fromtransmission of data in transmitter 210-p of transceiver 255-k (see FIG.2B). The output signal from NEXT/Echo interference filter 250-0 is inputto filter 574. Filter 574 can include digital filtering and alsoperforms an equalization.

Filter 574 equalizes the signal for intersymbol interference. Filter 574can include a feed-forward section, a feed-back section, or acombination of feed-forward and feed-back sections. The output signalfrom filter 574 can then be input to data recovery 575. Data recovery575 recovers the digital signal from the signals received from equalizerfilter 574. In some embodiments, data recovery 575 is a PAM slicer. Insome embodiments, data recovery 575 can also include an error correctiondecoder such as a trellis decoder, a Reed-Solomon decoder or otherdecoder. The output signal from data recovery 575 is then input todescrambler 576 so that the transmitted parallel bits are recovered.

Further examples and details of aspects of the above describedmulti-channel transceivers are described in U.S. application Ser. No.10/310,255, “Multi-Channel Communications Transceiver”, filed on Dec. 4,2002, by Sreen A. Raghavan, Thulasinath G. Manickam, Peter J. Sallaway,and Gerard E. Tayler, U.S. application Ser. No. 10/167,158,“Multi-Channel Communications Transceiver”, filed on Jun. 10, 2002, bySreen A. Raghavan, Thulasinath G. Manickam, Peter J. Sallaway, andGerard E. Tayler; U.S. application Ser. No. 10/071,771 to Sreen A.Raghavan, Thulasinath G. Manickam, Peter J. Sallaway, and Gerard E.Taylor; U.S. application Ser. No. 09/965,242 to Sreen Raghavan,Thulasinath G. Manickam, and Peter J. Sallaway, filed Sep. 26, 2001; andU.S. application Ser. No. 09/904,432, by Sreen Raghavan, filed on Jul.11, 2001, each of which is assigned to the same entity as is the presentapplication, each of which is herein incorporated by reference in itsentirety.

In some embodiments of the invention, NEXT/echo filter 250 may beimplemented ahead of cross-channel interference filter 570 and FEXTfilter 251 in the data stream. By implementing Next/echo filter 250first, the size of NEXT/Echo filter 250 may be smaller and/or have lesstaps. FEXT filter 251 and cross-channel interference filter 570 can movethe NEXT/Echo interference from one channel to another, thus it ispossible (or even likely) that a transform function. FN_(k) _(t) _(,k)_(r) ^(t,r) which may require few or no taps if cancellation of the NEXTand Echo interference was done first would need more taps aftercross-channel interference filter 570 and FEXT filter 251 have beenimplemented. The opposite phenomena, i.e. implementation of FEXT filter251 causing the FEXT interference to be transferred between channels,does not occur because NEXT/Echo filter 250 cancel utilizing transmitsymbols instead of received signals.

One skilled in the art will recognize that various components ofreceiver 272-r shown in FIG. 5 can be implemented in different ordersthan are shown. Further, although components are shown as circuitelements in this disclosure, it is understood that some if not all ofthe functions can be executed utilizing one or more digital processorsexecuting software code.

The embodiments of the invention described above are exemplary only andare not intended to be limiting. One skilled in the art will recognizevarious modifications to the embodiments disclosed that are intended tobe within the scope and spirit of the present disclosure. As such, theinvention is limited only by the following claims.

1. A transceiver, comprising: a) a receiver portion including at leastone receiver to receive signals from a complementary transmitter througha transmission medium, the at least one receiver including a pluralityof demodulators to receive the signals from a corresponding plurality offrequency separated channels: and b) an interference filter coupled tothe receiver portion to substantially reduce interference in signalsreceived by the receiver portion that result from transmission coupledto each of the plurality of corresponding frequency separated channelsfrom transmitters other than the complementary transmitter wherein theinterference filter includes a far-end cross-talk filter and the far-endcross-talk filter receives input signals from each of the plurality ofdemodulators of each of the at least one receiver and removesinterference from each of the signals in the receiver portion bysubtracting a selected portion of each of the input signals from each ofthe signals in the receiver portion.
 2. The transceiver of claim 1,wherein the far-end cross-talk filter further corrects for cross-channelinterference by subtracting from each of the signals in the receiverportion a selected contribution from each of the other signals in thereceiver portion.
 3. A transceiver, comprising: a) a receiver portionincluding at least one receiver to receive signals from a complementarytransmitter through a transmission medium, the at least one receiverincluding a plurality of demodulators to receive the signals from acorresponding plurality of frequency separated channels; and b) aninterference filter coupled to the receiver portion to substantiallyreduce interference in signals received by the receiver portion thatresult from transmission coupled to each of the plurality ofcorresponding frequency separated channels from transmitters other thanthe complementary transmitter wherein the transceiver further includesat least one transmitter and the interference filter includes a near-endcross talk and echo filter and the near-end cross talk and echo filtersubtracts from each of the signals received by the plurality ofdemodulators of each of the at least one receiver a selected portion oftransmit signals transmitted by the at least one transmitter.
 4. Thetransceiver of claim 3, wherein the at least one transmitter includes aplurality of modulators, each of the plurality of modulatorstransmitting into the corresponding plurality of frequency separatedchannels.
 5. The transceiver of claim 3, wherein the selected portion isadaptively chosen.
 6. A transceiver, comprising: a) a receiver portionincluding at least one receiver to receive signals from a complementarytransmitter through a transmission medium. the at least one receiverincluding a plurality of demodulators to receive the signals from acorresponding plurality of frequency separated channels wherein theplurality of demodulators includes at least one down-convertingdemodulator, the down-converting demodulator comprising: i) a downconverter coupled to receiver an analog signal from the transmissionmedium and generate base-band signals corresponding to signalstransmitted at a specific carrier frequency on the transmission medium;ii) an analog-to-digital converter coupled to the down converter toprovide a digitized base-band signal; and iii) a decoder coupled to theanalog-to-digital converter to provide data signals in response to thebase-band signals; and b) an interference filter coupled to the receiverportion to substantially reduce interference in signals received by thereceiver portion that result from transmission coupled to each of theplurality of corresponding frequency separated channels fromtransmitters other than the complementary transmitter.
 7. Thetransceiver of claim 6, further including an analog filter coupledbetween the down converter and the analog-to-digital converter, theanalog filter providing initial filtering and amplification to thebase-band signals.
 8. The transceiver of claim 7, wherein the analogfilter includes an amplifier.
 9. The transceiver of claim 6, wherein theanalog filter includes a low pass filter.
 10. The transceiver of claim6, wherein the decoder includes a trellis decoder.
 11. The transceiverof claim 10, wherein the trellis decoder decodes symbols from a QAMsymbol set.
 12. The transceiver of claim 6, wherein the plurality ofdemodulators includes a base-band demodulator, the base-band demodulatorcomprising: a) an analog processor coupled to receive the analog signalfrom the transmission medium; b) an analog-to-digital filter coupled tothe analog processor, the analog-to-digital filter providing a digitizedsignal; c) an equalizing filter coupled to the analog-to-digital filter;and d) a data-recovery block coupled to receive signals from theequalizing filter.
 13. The transceiver of claim 12, wherein the analogprocessor includes a low pass filter.
 14. The transceiver of claim 12,wherein the equalizing filter includes an equalizer.
 15. The transceiverof claim 12, wherein the data recovery block decodes data bits from aPAM symbol set.